Novel pulse transformer

ABSTRACT

A voltage converter comprises a second controller as a power switch of the secondary side of the transformer for comparing a detection voltage representing an output voltage and/or load current with a first reference voltage and generating a control signal, and a coupling element for transmitting the control signal generated by the second controller to the first controller on the primary side of the transformer enabling the first controller to generate a first pulse signal driving the power switch to control the on/off state of the primary side winding.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application is a Divisional Application of a pending U.S.patent application Ser. No. 15/240,879 filed on Aug. 18, 2016. TheDisclosure made in U.S. patent application Ser. No. 15/240,879 is herebyincorporated by reference.

U.S. patent application Ser. No. 15/240,879 claims the priority benefitof Chinese patent application number 201510579358.9 filed Sep. 11, 2015by a common inventor of this Application. The entire Disclosure made inthe Chinese patent application number 201510579358.9 is herebyincorporated by reference.

U.S. patent application Ser. No. 15/240,879 is a Continuation-In-Part(CIP) application of a pending U.S. patent application Ser. No.14/562,727 filed on Dec. 7, 2014. The entire disclosure made in U.S.patent application Ser. No. 14/562,727 is hereby incorporated byreference.

U.S. patent application Ser. No. 15/240,879 is a Continuation-In-Part(CIP) application of a U.S. patent application Ser. No. 14/562,729 filedon Dec. 7, 2014 and issued as U.S. Pat. No. 9,954,455 on Apr. 24, 2018.The entire disclosure made in U.S. patent application Ser. No.14/562,729 is hereby incorporated by reference.

U.S. patent application Ser. No. 15/240,879 is a Continuation-In-Part(CIP) application of a U.S. patent application Ser. No. 14/562,731 filedon Dec. 7, 2014 and issued as U.S. Pat. No. 9,577,542 on Feb. 21, 2017.The entire disclosure made in U.S. patent application Ser. No.14/562,731 is hereby incorporated by reference.

U.S. patent application Ser. No. 15/240,879 is a Continuation-In-Part(CIP) application of a U.S. patent application Ser. No. 14/562,733 filedon Dec. 7, 2014 and issued as U.S. Pat. No. 9,548,667 on Jan. 17, 2017.The entire disclosure made in U.S. patent application Ser. No.14/562,733 is hereby incorporated by reference.

U.S. patent application Ser. No. 15/240,879 is a Continuation-In-Part(CIP) application of a U.S. patent application Ser. No. 14/562,735 filedon Dec. 7, 2014 and issued as U.S. Pat. No. 9,577,543 on Feb. 21, 2017.The entire disclosure made in U.S. patent application Ser. No.14/562,735 is hereby incorporated by reference.

FIELD OF THE INVENTION

The invention mainly relates to an electronic device for voltageconversion, and in particular relates to a power supply device, which isused for sensing an output voltage or output current of secondarywindings of a transformer for power conversion in real time so as togenerate control signals with transient response and transmitting thecontrol signals to primary windings of the transformer for powerconversion by using coupling elements to control the primary windings tobe turned off or turned on.

BACKGROUND OF THE INVENTION

In a voltage converter, such as a pulse width modulation mode or pulsefrequency modulation mode converter, the voltage or the current of aload is acquired and a feedback signal representing the voltage or thecurrent of the load is fed back to a driving component of the voltageconverter via a feedback network. The duty ratio of a master switch,which is turned on and off in the voltage converter, is determinedthrough the driving component according to the feedback signal, so thatthe output voltage of the voltage converter at the load can be measured.It is known to a person having ordinary skill in the art that thedriving component of the voltage converter is used for driving themaster switch. However, the load voltage, which varies with time, cannotbe directly acquired from the load; the load voltage is instead sensedthrough the feedback network, which delays the load voltage measurementand thereby preventing synchronization of the driving component and thechange state of the load voltage to switch the master switch in realtime generating a difference between a present output voltage outputtedto the load and a practical voltage requested by the load, and thuscausing a potential instability for the output voltage.

BRIEF DESCRIPTION OF THE DRAWINGS

The characteristics and advantages of the invention will be apparentafter reading the following detailed descriptions and referring to thefollowing drawings.

FIG. 1 is a circuit diagram of a standard voltage converter.

FIG. 2 is a circuit diagram of a conventional feedback network for thevoltage converter.

FIG. 3 and FIG. 4 are the circuit diagrams of voltage converters withthe coupling elements including capacitors or pulse transformersrespectively.

FIG. 5 is a circuit diagram of a starting module equipped in a firstdrive on the primary winding of the voltage converter.

FIG. 6A is a circuit diagram showing the mode that a control signal istransmitted to the first drive from a second controller of the secondarywinding by using a capacitance coupling element.

FIG. 6B are waveforms showing a first pulse signal and a second pulsesignal generated along with change of the output voltage or current onthe basis of FIG. 6A.

FIG. 6C is a circuit diagram showing the mode for enabling the turn-ontime of the master switch to be adjustable in the second controller onthe basis of FIG. 6A.

FIG. 6D is waveforms showing the adjusting the turn-on time on the basisof FIG. 6C.

FIG. 7A is a circuit diagram showing the mode that the control signal istransmitted to the first drive from the second controller of thesecondary winding by using the pulse transformer.

FIG. 7B are waveforms showing that the first pulse signal and the secondpulse signal are generated along with change of the output voltage orcurrent on the basis of FIG. 7A.

FIG. 7C is a circuit diagram showing that output results of a filter andan amplifier are overlapped and are further compared with a referencevoltage on the basis of FIG. 7A.

FIG. 8 is a circuit diagram of a voltage converter showing that asynchronous switch of the secondary winding is replaced by a rectifierdiode of the secondary winding.

FIG. 9 is a circuit diagram showing the mode for adjusting the turn-ontime of the master switch when the load is lightened.

FIG. 10 is waveforms showing that the master switch turn-on timedetermined by a later control signal is suppressed by a former controlsignal on the basis of FIG. 9.

FIGS. 11A-11B are schematic diagrams illustrating the structure of apulse transformer according to a first embodiment.

FIGS. 12A-12E are schematic diagrams illustrating the structure of apulse transformer according to a second embodiment.

FIGS. 13A-13C are schematic diagrams illustrating the structure of apulse transformer according to a third embodiment.

DETAILED DESCRIPTION OF THE EMBODIMENTS

With the combination of the embodiments, the technical scheme of theinvention is clearly and completely illustrated, the describedembodiments are only embodiments for describing the invention but notall embodiments, based on the embodiments, schemes obtained bytechnicians of the field without creative work all belong to theprotection scope of the invention.

As shown in FIG. 1, an AC/DC FLYBACK voltage converter includes a powertransformer T for voltage conversion mainly including primary windingsL_(P) and secondary windings L_(S), where the first end of the primarywinding L_(P) is used for receiving an input voltage V_(IN) at an inputnode N₁₀, and a master switch Q1 is connected between a second end ofthe primary winding L_(P) and a ground terminal GND. The basic workingmechanism is that the master switch Q1 is driven to be turned on andturned off through a primary winding controller, which is also referredto as a first controller 104. When the master switch Q1 is turned on,the current of the primary windings flows through the primary windingsL_(P) and the master switch Q1 and to the ground terminal GND, thereforeno current flows through the secondary windings L_(S) in this period,and thus the primary windings L_(P) start to store energy. Once themaster switch Q1 is turned off, the current of the primary windingsL_(P) is stopped, thus the polarities of all windings are reversed, andthe transformer T starts to transfer the energy to the secondarywindings L_(S), so that the secondary windings Ls are enabled to providethe working voltage and current to the load 18 while the master switchQ1 is turned off. An output capacitor C_(OUT) is charged at the outputnode N₂₀, and the working voltage can be continuously provided to theload 18 through the output capacitor C_(OUT) when the working currentcannot be directly provided to the load 18 as no current flows throughthe secondary windings L_(S). In some embodiments, the transformer Tfurther comprises an auxiliary winding L_(AUX), where the coils of theauxiliary winding L_(AUX) are wound in a direction same as those of thesecondary windings L_(S), once the master switch Q1 is turned off, thecurrent flowing through the auxiliary winding L_(AUX) can be used forcharging a capacitor C_(AUX) and can be taken as a working voltagesource of a first controller 104.

The alternating current is firstly rectified by using a bridge rectifier101 comprising four diodes D11 to D14. Generally, a sinusoidalalternating current voltage V_(AC) is inputted into a pair of inputlines, for example buses 12 and 14, and the bridge rectifier 101 makesfull use of the positive semi-cycle and the negative semi-cycle of thesinusoidal waveforms of an original alternating current to convertcomplete sinusoidal waveforms of the alternating current of samepolarity to output. After full-wave rectification of the bridgerectifier 101, alternating current V_(AC) is converted into a pulsatingvoltage with the alternating current. For further reducing the waves ofthe pulsating voltage, a CLC filter L₁ C₁₁ C₁₂ is adopted to filter outthe waves of the rectified voltage so as to obtain an input voltageV_(IN) after the alternating current is rectified. As shown in FIG. 1, afirst end of an inductor L₁ of the CLC filter is connected with thecathodes of diodes D₁₁ and D₁₃ of the rectifier 101, while a second endof the inductor L₁ is coupled to the first end of the primary windingL_(P) at a node N₁₀. Capacitor C₁₁ of the CLC filter is connectedbetween the first end of the inductor L₁ and the ground terminal GNDwhile another capacitor C₁₂ of the CLC filter is connected between thesecond end of the inductor L₁ and the ground terminal GND. The anodes ofthe diodes D₁₂ and D₁₄ of the bridge rectifier are both connected to theground terminal GND, while the bus 12 is connected with the anode of thediode D₁₁ and the cathode of the diode D₁₂, and the bus 14 is connectedwith the anode of the diode D₁₃ and the cathode of the diode D₁₄.

The voltage converter further comprises an RCD clamping circuit or aturn-off buffer circuit 103 which is connected in parallel with theprimary windings L_(P). The turn-off buffer circuit 103 comprises acapacitor and a resistor, which are connected in parallel with eachother and mutually connected with the node N₁₀ at one end of thecapacitor and the resistor, and connected to the cathode of a diode inthe turn-off buffer circuit 103 at the other end. The anode of the diodeis connected with the second end of the primary windings L_(P). Theturn-off buffer circuit 103 limits the overlap of a peak voltage and aprimary coil reflection voltage caused by energy of high-frequency valueconverter leakage inductance when the master switch Q1 is turned off.Typically, an overlap voltage may be generated when the master switch Q1is turned off from a saturated state; thus the energy of leakageinductance can be adopted to charge the capacitor through the diode ofthe turn-off buffer circuit 103. The voltage of the capacitor can beincreased to the overlap value of counter electromotive force and theleakage inductance voltage, and the capacitor has the function ofabsorbing the energy. When the primary windings L_(P) and the masterswitch Q1 enter into the turn-on period from the turn-off state, theenergy of the capacitor of the turn-off buffer circuit 103 is releasedthrough the resistor of the turn-off buffer circuit 103 until thevoltage of the capacitor meets the counter electromotive force beforethe master switch Q1 is turned off at the latter time.

The first end of the secondary winding Ls is connected with the outputnode N₂₀, the second end of the secondary winding L_(S) is connectedwith the first end of the synchronous switch Q2, and the second end ofthe synchronous switch Q2 is connected with the reference groundpotential VSS. An output capacitor C_(OUT) is connected between theoutput node N₂₀ and a reference ground potential VSS, an output voltageV_(O), or the working voltage of load 18, can be provided to the load 18at the output node N₂₀. It is required that if one of the two switchesQ1 and Q2 is turned on, the other one needs to be turned off. Forexample, the synchronous switch Q2 of the secondary winding is turnedoff if the master switch Q1 of the primary winding is turned on; viceversa, the synchronous switch Q2 of the secondary winding is turned onif the master switch Q1 of the primary winding is turned off. The masterswitch Q1 and the synchronous switch Q2 are respectively provided with afirst end, a second end and a control end; whether the first ends andthe second ends of the switches are in communication is determinedaccording to logic state (i.e., high or low) of the signals applied tothe control ends. In the normal working period of the voltage converter,a first pulse signal S₁ generated by the first controller 104 of theprimary winding is adapted to drive the master switch Q1, by turning itoff and on. A second pulse signal S₂ generated by a second controller105 of the secondary winding is adapted to drive the synchronous switchQ2, by turning it off and on. In addition, when the synchronous switchQ2 is driven by the second pulse signal S₂ generated by the secondcontroller 105, a dead time between the master switch Q1 and thesynchronous switch Q2 can be generated causing that the synchronousswitch Q2 is turned off under the control of the second pulse signal S₂while the master switch Q1 is also turned off under the control of thefirst pulse signal S₁.

In addition to the secondary windings L_(S), the first end of theauxiliary winding L_(AUX) is connected with the anode of a diodeD_(AUX), while the cathode of the diode D_(AUX) is connected with afirst end of the capacitor C_(AUX). The second end of the auxiliarywinding L_(AUX) and the second end of the capacitor C_(AUX) is connectedwith the ground terminal GND. When the master switch Q1 is turned on,the first ends of the secondary windings L_(S) and the auxiliary windingL_(AUX) are negative relative to their second ends and have no currentflowing through the windings; the load 18 is supplied with power of theoutput capacitor C_(OUT). Conversely, when the master switch Q1 isturned off, the secondary windings L_(S) and the auxiliary windingL_(AUX) are of opposite polarities; the respective first ends arepositive relative to the second ends and both have current flowingthrough the windings. Thus, the energy from the primary windings L_(P)is transferred to the secondary windings L_(S) and the auxiliary windingL_(AUX). In other words, when the master switch Q1 is turned off, thesecondary windings L_(S) not only provide current to the load 18, butalso charge the output capacitor C_(OUT), and the auxiliary windingL_(AUX) also charges the auxiliary capacitor C_(AUX) as a power supply.As shown in FIG. 1, the voltage V_(CC) held at one end of the auxiliarycapacitor C_(AUX) is the power supply voltage of the first controller104. A safety capacitor C_(Y), which is connected between the groundterminals GND of the primary windings and the reference groundpotentials VSS of the secondary windings, by which the noise voltagegenerated by the capacitor between the primary windings and thesecondary windings may be filtered out, or common mode interferencecaused by a coupling capacitor between the primary windings and thesecondary windings can be filtered out.

The second controller 105 of the secondary winding is adapted to capturethe change of the output voltage V_(O) at the node N₂₀ in real time, orsense the change of the load current I_(O) (i.e. output current) flowingthrough the load 18 in real time, and thus a control signal SQ isgenerated. A first pulse signal S1 can be further generated by the firstcontroller 104 of the primary winding according to the high/low logicstates of the control signal SQ, and thus whether the master switch Q1needs to be turned on or turned off can be determined according to thefirst pulse signal S1. As the control signal SQ generated by the secondcontroller 105 changes nearly in a temporary state response mannerrelative to the voltage V_(O) or current I_(O), the first pulse signalS1 generated by the first controller 104 can respond to the change ofthe control signal SQ in real time. Equivalently, the first pulse signalS1 tracks the change of the voltage V_(O) or current I_(O) in real time.The detail of how control signal SQ is generated by the secondcontroller 105 and how the information is transmitted between the secondcontroller 105 and the first controller 104 through the coupling elementis described in detail below.

As shown in FIG. 2, a conventional feedback network includes a resistorR₁ and a resistor R₂ adapted for partial voltage sampling on the outputvoltage V_(O); a resistor R₃ adapted for loop gain adjustment; andcompensation capacitors C₁ and C₂, and a compensation resistor R₅adapted for compensation. The general working principle of the feedbacknetwork is that when the output voltage V_(O) is increased, the partialvoltages of the resistors R₁ and R₂ are inputted into a control end(i.e., an input end of a voltage error amplifier) of a three-endprogrammable in-parallel voltage stabilizing diode in the feedbacknetwork, so that the voltage of the control end is increased along withincrease of the output voltage V_(O). As the voltage of the cathode(i.e. an output end of the voltage error amplifier) of the three-endprogrammable in-parallel voltage stabilizing diode drops, a primary sidecurrent I_(P), which flows through a light emitting element connectedbetween the cathode of the three-end programmable in-parallel voltagestabilizing diode and the resistor R₃ in an optical coupler 17, isincreased; an output current flowing through a transistor for receivinglight intensity on another side of the optical coupler 17 is alsoincreased. When the voltage of a feedback port COMP of a primary windingcontroller 16 is reduced, the duty ratio of the pulse signal forcontrolling the master switch Q1 is also reduced, and consequently theoutput voltage V_(O) is also reduced. Conversely, when the outputvoltage V_(O) is reduced, the adjustment process is similar but inopposite tendency—increase in the duty ratio of the pulse signal forcontrolling the master switch Q1 also causes the voltage V_(O) toincrease. The resistor R4 provides additional currents into the feedbacknetwork to prevent it from operating abnormally when the current is toosmall. The resistor R₄ can be omitted if the resistor R3 has anappropriate resistance. The feedback network shown in FIG. 2 requiressufficient gain and phase margin to ensure the stability of the wholesystem. For example, the open loop gain at least needs a phase margin of45 degrees, and generally, the phase margins ranges from 45 degrees to75 degrees. However, in the conventional feedback network of FIG. 2, thecontrol mode is complex and the delay effect is conspicuous, as such thesituation of the secondary winding cannot be detected by the primarywinding controller 16 in real time.

As shown in FIG. 3, the coupling element 106 of the voltage convertershown in FIG. 1 includes a coupling capacitor. Alternatively, as shownin FIG. 4, the coupling element 106 of the voltage converter shown inFIG. 1 includes a pulse transformer. In addition, the coupling element106 can include other dielectric elements or optical coupling elementsas long as data information can be interacted between the primarywinding controller that is also called as the first controller 104 andthe secondary winding controller that is also called as the secondcontroller 105.

As shown in FIG. 5, a safety capacitor C_(X), which is used forsuppressing different model interference and filtering outhigh-frequency clutter signals, is connected between the input lines 12and 14, and one input capacitor C_(IN) is connected between the inputnode and the ground terminal GND. The alternating current voltage V_(AC)inputted into the input lines 12 and 14 is rectified by the bridgerectifier 101 and is subsequently filtered by the input capacitorC_(IN), so as to obtain the input voltage V_(IN). The voltage converterconverts the input voltage V_(IN) to provide the output voltage V_(O) tothe load through output lines 22 and 24. In this embodiment, the deviceof the invention further comprises a rectifier circuit connected withthe input lines 12 and 14. The rectifier circuit includes a rectifyingdiode D₂₁ having the anode connected with the input line 12 and anotherrectifying diode D₂₂ having the anode connected with the input line 14;the cathodes of the diodes D₂₁ and D₂₂ are both connected with the drainof a high-voltage starting element JFET (junction field effecttransistor) of the first controller 104. The limiting resistor R₂₁ shownin FIG. 1 may also be connected between the drain of JFET and thecathodes of the diodes D₂₁ and D₂₂. The source of JFET is connected withthe anode of a diode D₃₁, and the cathode of the diode D₃₁ is connectedwith one end of the auxiliary capacitor C_(AUX), which is connected withthe ground and used as the power supply. A limiting resistor R₃₁ isconnected between a gate control end and the source of the JFET. Acontrol switch SW₃₁ is connected between the gate of the JFET and theground terminal GND. The first end of the control switch SW₃₁ isconnected with the gate of the JFET, and the second end of the controlswitch SW₃₁ is connected with the ground terminal GND. When the inputlines 12 and 14 are supplied with the alternating currents, an on-offsignal CTRL applied to the gate of the control switch SW₃₁ starts todrive the control switch SW₃₁ to enter into a turn-on state. The gate ofthe control switch SW₃₁ can be connected with the ground potential GNDto communicate with JFET of a negative critical voltage, so that thegenerated current flows from the drain to the source to charge one endnot connected with the ground of the capacitor C_(AUX) through the diodeD₃₁. Forward voltage drop across resistor R₃₁ is increased, but thevoltage between the gate and the source is decreased, so that thevoltage between the source and the gate of the JFET is approximatelybalanced with a voltage of Pinch-off of the JFET. Specifically, theactual voltage drop from the gate G to the source S of the JFET is equalto a negative value of the Pinch-off voltage. When the capacitor C_(AUX)is charged by the JFET until the stored voltage V_(CC) is increased tomeet a starting voltage, a driving control module (not shown), which isadapted to generate an initial pulse signal, may be triggered to enterinto a working state. The master switch Q1 is driven by the initialpulse signal to be turned on or off. The above steps complete theStart-Up procedure for the voltage converter. After the Start-Upprocedure is completed, the capacitor C_(Aux) is charged through thediode D_(AUX) by using an auxiliary winding L_(AUX). In addition, avoltage divider can be used to connect between the first end of theauxiliary winding L_(AUX) and the ground terminal GND. The partialvoltage sampled by the voltage divider can be inputted into the firstcontroller 104, so that zero current passage (ZCD) detection on thesecondary windings or over-voltage detection on the output voltages ofthe secondary windings can be achieved by using the voltage dividerthrough the first controller 104 (not shown). As shown in FIG. 1, thefirst end of the master switch Q1 (i.e., the drain) is connected withthe second end of each primary winding L_(P), a sensing resistor R_(S)is further connected between the second end, which is the source of themaster switch Q1, and the ground terminal GND, thus the voltage V_(S) ofthe current flowing through the primary windings can be obtained bymultiplying the current flowing through the primary windings L_(P) withthe resistance of the sensing resistor R_(S), and if the voltage V_(S)is inputted into the first controller 104 and is defined during a presetlimiting voltage V_(LIMIT) by the first controller 104, the currents ofthe primary windings can be monitored, and over-current protection canbe achieved.

As shown in FIG. 1, after the starting procedure is complete and themaster switch Q1 is toggled for the first time, the voltage captured atthe first end of the secondary winding L_(S) is used as the startingvoltage ST to start the second controller 105 of the secondary winding.The second controller 105 is adapted to monitor the output voltage V_(O)of the secondary winding and the current I_(O) flowing through the load18 in real time. Specifically, a partial voltage V_(FB) representing theoutput voltage V_(O) is captured by the voltage divider comprising apair of serially connected resistors R_(D1) and R_(D2) between theoutput node N₂₀ and the reference ground potential VSS of the secondarywinding. Specifically, V_(FB) is measured at a joint node of theresistor R_(D1) and the resistor R_(D2). V_(FB) is then used as afeedback voltage and inputted into the second controller 105. The load18 and a sensing resistor R_(C) are serially connected and arrangedbetween the output node N₂₀ and the reference ground potential VSS ofthe secondary winding. The value of the current I_(O) flowing throughthe load 18 obtained by dividing the sensing voltage drop V_(CS) of thesensing resistor R_(C) by the resistance of the sensing resistor R_(C).In other words, the sensing voltage drop V_(CS) can be used to representthe loading current flowing through the load 18 and the sensing resistorR_(C).

FIG. 6A illustrates the components of the first controller 104 and thesecond controller 105 used for controlling the turn on/turn off of themaster switch Q1 in real time according to the change of the sensingvoltage drop V_(CS) and the feedback voltage V_(FB) mentioned above.Data interaction of the first controller 104 and the second controller105 is implemented through the coupling element 106, which comprises twocoupling capacitors C₂₁ and C₂₂. The working mechanisms of the firstcontroller 104 and the second controller 105 are described in detailbelow. The structures of the first controller 104 and the secondcontroller 105 shown in FIG. 6A are only an example according to anembodiment of the present invention, thus other equivalenttransformation modes and schemes obtained on the basis of thisembodiment also belong to the protection scope of the invention.

The second controller 105 comprises a first switch SW₄₁ and a secondswitch SW₄₂, each of which includes a first end, a second end and acontrol end. Whether the first end and the second end are incommunication is determined according to the high/low logic states ofsignals applied by the control ends. The first switch SW₄₁ and thesecond switch SW₄₂ are serially connected between a bias circuit 105 dand the reference ground potential VSS. For example, the first end ofthe first switch SW₄₁ is connected with the bias circuit 105 d; thesecond end of the first switch SW₄₁ is connected with the first end ofthe second switch SW₄₂; and the second end of the second switch SW₄₂ isconnected with the referential ground potential VSS. The first switchSW41 and the second switch SW₄₂ are controlled by a control signal SQgenerated by the output end Q of the RS trigger 105 a (a port Q of theRS trigger is defined as an output end; a port QN is defined as anon-end Q or a complementary output end). For example, the controlsignal SQ is coupled with the control end of the first switch SW₄₁ afterpassing through a buffer and coupled with the control end of the secondswitch SW₄₂ through an inverse phase signal generated by a phaseinverter 105 e. As such, when-n the first switch SW₄₁ is turned on, thesecond switch SW₄₂ needs to be turned off, or when the first switch SW₄₁is turned off, the second switch SW₄₁ needs to be turned on.

The resistor R_(D1) and the resistor R_(D2) of the voltage divider(FIG. 1) divide and capture a partial voltage of the output voltageV_(O), which is the feedback voltage V_(FB). The feedback voltage V_(FB)is inputted into an inverting input terminal of a first comparator A1 inthe second controller 105, while a first reference voltage V_(REF) isinputted into a non-inverting input end of a first comparator A1. Inanother embodiment, the sensing resistor R_(C) serially connected withthe load 18 captures the sensing voltage V_(CS) flowing through the load18, and the sensing voltage V_(CS) is inputted into the inverting inputterminal of the first comparator A1 in the second controller 105. Theoutput end of the first comparator A1 is connected with a setting end Sof the RS trigger 105 a. A signal S_(ON) outputted from an on-timegenerator 105 c in the second controller 105 is inputted into a resetend R of the RS trigger 105 a, and a one-shot trigger 105 b is connectedbetween the output end Q of the RS trigger 105 a and the on-timegenerator 105 c. In the circuit from the first switch SW₄₁ and thesecond switch SW₄₂ to the reference ground potential VSS in the secondcontroller 105, a node N₂ serves as a common node for the second end ofthe first switch SW₄₁ and the first end of the second switch SW₄₂, and anode N₄ is at the second end of the second switch SW₄₂ and connectedwith the reference ground potential VSS.

The first controller 104 comprises a second comparator A2, a node N₁connected with the non-inverting input terminal of the second comparatorA2, a node N₃ connected with the ground terminal GND, and a resistor R₄₁connected between the nodes N₁ and N₃. A second reference voltage V_(TH)is inputted into the inverting input terminal of the second comparatorA2. A capacitor C₂₁ of the coupling element 106 is connected between thenode N₁ of the first controller 104 and the node N₂ of the secondcontroller 105, and a capacitor C₂₂ of the coupling element 106 isconnected between the node N₃ of the first controller 104 and the nodeN₄ of the second controller 105. The coupling element 106 has similardata transmission effects as an Ethernet. For example, the node N₁ canbe taken as a receiving interface RX1+ of the first controller 104, thenode N₃ can be taken as a receiving interface RX2− of the firstcontroller 104, correspondingly, the node N₂ can be taken as atransmitting interface TX1+ of the second controller 105, and the nodeN₄ can be taken as a transmitting interface TX2− of the secondcontroller 105.

A first pulse signal S₁ for controlling the master switch Q1 isgenerated through the cooperation of the first controller 104 and thesecond controller 105 as shown in FIGS. 6A and 6B. When the feedbackvoltage V_(FB) or the sensing voltage V_(CS) is inputted into theinverting end of the first comparator A1 in the second controller 105,and when the feedback voltage V_(FB) or the sensing voltage V_(CS)starts to be lower than the first reference voltage V_(REF) inputted atthe non-inverting end, which occurs at the moment T₁ in FIG. 6B, theoutput result of the first comparator A1 is at a logic high level, sothat the RS trigger 105 a outputs the control signal SQ from the outputend Q at the logic high level. Thus, the control signal SQ iscommunicated with the first switch SW₄₁ in FIG. 6A, and the secondswitch SW₄₂ is turned off as the control signal SQ is at the logic lowlevel after passing the phase inverter 105 e. As the second switch SW₄₂is turned off when the first switch SW₄₁ is turned on, the referenceground potential VSS is lower than the potential of the ground terminalGND, signal is transmitted between the second controller 105 and thefirst controller 104 forming a current channel on a LOOP1 comprising thebias circuit 105 d, the first switch SW₄₁, the node N₂, the capacitorC₂₁, the node N₁, the resistor R₄₁, the node N₃, the capacitor C₂₂, thenode N₄, and the reference ground potential VSS, as such a positivepower supply source provided by the bias circuit 105 d flows through thefirst switch SW₄₁ and the node N₂ and starts to charge the capacitor C₂₁in the coupling element 106 changing the charge voltage V_(TX1) at thenode N₂, or the transmitting interface TX1+, as shown in FIG. 6B withthe charge voltage V_(TX1) increased gradually. The change of the chargevoltage V_(RX1) at the node N1, or the receiving interface RX1+, is alsoshown in FIG. 6B. As the voltages at two ends of the capacitor C₂₁ arenot mutated, the maximum value of the voltage V_(RX1) is achieved at themoment T1, and the voltage V_(RX1) at the receiving interface RX1+ isgradually reduced while the voltage of a polar plate of the capacitorC₂₁ is gradually increased. In the period from T₁ to T₂, as the chargevoltage V_(RX1) at the node N₁, or the receiving interface RX1+, isgreater than the second reference voltage V_(TH), the first pulse signalS₁ outputted from the second comparator A2 is at the logic high leveland is coupled with the control end of the master switch Q1. As thefirst pulse signal Si already starts to control the master switch Q1, inthe Start-Up period of the voltage converter, the initial pulse signaloutputted from the driving control circuit and used for controlling themaster switch Q1 in the first controller 104 is stopped, so that themaster switch Q1 is completely controlled by the first pulse signal S₁unless the master switch Q1 needs to be started by the initial pulsesignal to start the voltage converter.

As shown in FIG. 6B, the first pulse signal S₁ extends from the momentT₁ to a moment T₂, then the turn-on time T_(ON) set by the on-timegenerator 105 c is ended. A signal S_(ON) at the logic high levelgenerated by on-time generator 105 c and used as a reset signal istransmitted to the reset end S of the RS trigger 105 a, thus the controlsignal SQ outputted from the output end Q of the RS trigger 105 a isconverted to the logic low level, which turns off the first switch SW₄₁in FIG. 6A. However, the second switch SW₄₂ is turned on when thecontrol signal SQ is inverted to the logic high level after passedthrough the phase inverter 105 e. As the second switch SW₄₂ is turnedoff when the first switch SW₄₁ is turned on, a part of charges stored inthe capacitor C₂₁ and the capacitor C₂₂ is consumed by the resistor R₄₁from the second controller 105 to the first controller 104 along aclosed LOOP2 comprising the node N₂, the second switch SW₄₂, the nodeN₄, the capacitor C₂₂, the node N₃, the resistor R₄₁, the node N₁, thecapacitor C₂₁ and the node N₂. Therefore, from the moment T₂, chargesare released from the capacitor C₂₁, then the charge voltage V_(TX1) atthe node N₂, or the transmitting interface TX1+, is gradually reduced.At the moment T₂, as the voltage of the capacitor C₂₁ is not mutated,the voltage V_(TX1) at the node N₁, or the receiving interface RX1+, canbe reduced to be temporarily negative; along with charge release of thecapacitor C₂₁ and the capacitor C₂₂, and the voltage V_(RX1) at thereceiving interface RX1+ is approximately equal to zero potential at amoment T₃. The voltage V_(TX1) at the node N₂, or the transmittinginterface TX1+, is also approximately equal to zero potential at themoment T₃. In the period from T₂ to T₃, as the voltage V_(RX1) at thenode N₁, or the receiving interface RX1+, is less than a secondreference voltage V_(TH), for example approximately to the zeropotential, the first pulse signal S₁ outputted from the secondcomparator A2, is at the logic low level, thus the master switch Q1 isturned off. As shown in FIG. 6B, the turn-on time T_(ON) from the momentT₁ to the moment T₂ is the period that the master switch Q1 is turned onand the turn-off time T_(OFF) from the moment T₂ and the moment T₃ isthe period that the master switch Q1 is turned off. In addition, asshown in FIG. 1, the second pulse signal S₂ is the inverse phase signalof the first pulse signal S₁ or the control signal SQ, so that the logicstates of the second pulse signal S₂ at the turn-on time T_(ON) and theturn-off time T_(OFF) are opposite to those of the first pulse signalS₁, and the second controller 105 is adapted to generate the secondpulse signal S₂ for controlling the synchronous switch Q2 of thesecondary winding.

In the period the master switch Q1 is turned on, primary current flowsthrough the primary winding L_(P) to store energy, and at the moment, asthe synchronous switch Q2 is turned off, no current flows through thesecondary winding L_(S), and power can be provided to the load 18through the output capacitor C_(OUT). In the period that the masterswitch Q1 is turned off, the primary current is reduced to zero, theenergy of the primary winding L_(P) is transferred to the secondarywinding L_(S) and the auxiliary winding L_(AUX), which turns on thesynchronous switch Q2, thus current flows through the secondary windingL_(S) and the synchronous switch Q2. The load 18 is provided withcurrent from the secondary winding L_(S) and the output capacitorC_(OUT) is charged, while the capacitor C_(AUX) is also charged withpower from the auxiliary winding L_(AUX). The time-delay measurement forthe turn-on time T_(ON) is determined by the on-time generator 105 c. Asshown in FIG. 6A and FIG. 6B, the one-shot trigger 105 b can betriggered at the rising-edge of the control signal SQ outputted from theRS trigger 105 a generating one temporary state pulse signal CLK1 of ananosecond grade (the pulse signal CLLK outputted from the one-shottrigger or the one-shot circuit is generally in two logic states oftemporary state and steady state). A narrow temporary state pulse signalCLK1 is at the high level (in the temporary state period) at the momentof rising-edge of the control signal SQ, and is at the low level atother moments (in the steady state period). The temporary state pulsesignal CLK1 at the high level is adapted to inform the on-time generator105 c to start to time, and a signal S_(ON) at the high level istransmitted from the on-time generator 105 c to reset the RS trigger 105a at the moment that the preset turn-on time T_(ON) is just reached,therefore, the control mode is a constant on time control mode, and inthe present invention, in each switch period, the constant on timeT_(ON) can be also adjusted, for example, a minimum constant on timeT_(ON-MIN) or maximum constant on time T_(ON-MAX) that meets therequirements can be designed.

FIG. 6C is an alternative mode of the one in FIG. 6A. The on-offfrequency f of the master switch Q1 is reduced as the input voltageV_(IN) is increased vice versa, and the frequency f is reduced as theturn-on time T_(ON) is increased or vice versa. If the on-off frequencyf is too small, the magnetic core flux of the transformer T cannot berecovered to the starting point of a hysteresis loop and a magnetic coreis over-saturated. The transformer T can be saturated if the on-offfrequency f is too small as the input voltage V_(IN) is increased, andat the moment the magnetic core can be easily burn if the voltage is notgenerated. In this embodiment, the problems can be overcome. When themaster switch Q1 is turned on and the synchronous switch Q2 is turnedoff, no current flows through the secondary winding L_(S), but thevoltage sampling V_(SAM) captured at the second end of the secondarywinding L_(S) and the first end of the synchronous switch Q2 at themoment is generally equal to the ratio of the number of turns NS of thesecondary winding L_(S) to the number of turns NP of the primary windingL_(P) multiplied with the input voltage V_(IN). In the other words, thevoltage V_(SAM) is associated with the input voltage V_(IN). The voltageV_(SAM) can be sensed by the on-time generator 105 c, and therefore anappropriate turn-on time T_(ON) is designed to inhibit magnetic coresaturation caused by abnormal state of the on-off frequency value f. Asshown in FIGS. 6C and 6D, if the sensing voltage drop V_(CS) or thefeedback voltage V_(FB) is less than the first reference voltageV_(REF), the first comparator A1 outputs a high level signal to thesetting end S of the RS trigger 105 a, and the control signal SQgenerated by the output end Q of the RS trigger 105 a is turned from thelow level to the high level, and the one-shot trigger 105 b generates ahigh level temporary state pulse signal CLK1 at the rising-edge as thecontrol signal SQ is turned from the low level to the high level whenthe control signal SQ is sent to the one-shot trigger 105 b. The on-timegenerator 105 c comprises a sampling holder (S/H) 105 c-1, avoltage-current converter 105 c-2, a third switch SW₅₁ and a capacitorC_(T). The input end of the sampling holder 105 c-1 is connected withthe second end of the secondary winding L_(S), while the output end ofthe sampling holder 105 c-1 is connected with the voltage input end ofthe voltage-current converter 105 c-2 provided with a working voltagethrough the power supply voltage V_(DD). The current output end of thevoltage-current converter 105 c-2 and one end of the capacitor C_(T) areconnected with a node N_(T), and another end of the capacitor C_(T) isconnected with the ground terminal GND. The first end of the thirdswitch SW₅₁ is connected with the node N_(T), and the second end isconnected with the ground terminal GND, so that the third switch SW₅₁and the capacitor C_(T) are connected in parallel, and the temporarystate pulse signal CLK1 generated by the one-shot trigger 105 b isinputted into the control end of the third switch SW₅₁. The on-timegenerator 105 c further comprises a third comparator A3 having thenon-inverting input terminal connected with the node N_(T) and theinverting end inputted with a third reference voltage V_(P).

As shown in FIG. 6C, the on-time T_(ON) is adjusted by the on-timegenerator 105 c such that the sampling holder 105 c-1 is provided asample voltage V_(SAM) from the second end of the secondary windingL_(S) when the master switch Q1 is turned on and the synchronous switchQ2 is turned off, such that the larger the input voltage V_(IN) is, thehigher the voltage V_(SAM) held by the sampling holder 105 c-1 is, thelarger the current outputted from the voltage-current converter 105 c-2is; as is vice versa. When the temporary state pulse signal CLK1 fordriving the third switch SW₅₁ is at the high level at the rising-edge ofthe control signal SQ generated by the RS trigger 105 a, which is at thelow level at other time, the third switch SW₅₁ is turned on at themoment of rising-edge of the control signal SQ, thus the charges storedat one end of the capacitor C_(T), i.e., at the node N_(T), arereleased; and then a signal S_(ON) at the low level is generated fromthe output end of the third comparator A3 at the moment. As shown inFIG. 6D, a preset time T_(SET) is started at the rising-edge of thecontrol signal SQ. After the rising-edge of the control signal SQ, thetemporary state pulse signal CLK1 is converted to the low level again;the third switch SW₅₁ is turned off and the capacitor C_(T) is chargedwith power through the current outputted from the voltage-currentconverter 105 c-2. After the capacitor C_(T) is charged in the on-timeperiod T_(ON) and after the on-time period T_(ON) is ended, the voltageat the node N_(T) is greater than the third reference voltage V_(P)during the off-time period T_(OFF). As a result, the signal S_(ON)generated at the output end of the third comparator A3 is turned to thehigh level during the off-time period T_(OFF) from the low level duringthe on-time period T_(ON) and then is inputted into the reset end R ofthe RS trigger 105 a to reset the RS trigger 105 a, and thus controlsignal SQ generated by the output end Q of the RS trigger 105 a ischanged to the low level during the off-time period T_(OFF) from thehigh level during the on-time period T_(ON). The control signal SQ iscontinuously at the low level until the off-time period T_(OFF) is endedunless the sensing voltage drop V_(CS) or the feedback voltage V_(FB) isless than the first reference voltage V_(REF), and the first comparatorA1 regenerates the high level signal to set the RS trigger 105 a tooutput the high level control signal SQ. The signal S_(ON) generated atthe output end of the third comparator A3 is continuously at the highlevel during the off-time period T_(OFF) until the off-time period isended unless the control signal SQ has a rising-edge, then the temporarypulse signal CLK1 is at the high level to turn on the third switch SW₅₁,the node N_(T) of the capacitor C_(T) is in transient discharge, and asignal S_(ON) at the low level is then generated by the third comparatorA3.

As mentioned above, the larger the input voltage V_(IN) is, the largerthe voltage held by the sampling holder 105 c-1 is, and then the largerthe current outputted from the voltage-current converter 105 c-2 is, sothat the charge time is shortened, thus the voltage at the node N_(T) atone end of the capacitor C_(T) can rapidly exceed the third referencevoltage V_(P), and the T_(ON) is shortened during the whole on-offperiod with the control signal SQ at the high level and the masterswitch Q1 turned on during the period T_(ON). As a result the larger theinput voltage V_(IN) is, the shorter the on-time T_(ON) is, andcorrespondingly, the control signal SQ during the period T_(OFF) is atthe low level and the master switch Q1 is turned off during this period.In other words, although the input voltage V_(IN) is increased, theon-off frequency value f is reduced, and the reduction of the on-offfrequency value f is suppressed consequently when the on-time T_(ON) isshortened. Vice versa, the smaller the input voltage V_(IN) is, thesmaller the voltage held by the sampling holder 105 c-1 is, and then thesmaller the current outputted from the voltage-current converter is, sothat the charge time can be prolonged and the voltage at the node N_(T)at one end of the capacitor C_(T) can exceed the third reference voltageV_(P) in a relatively slow speed, thus the period T_(ON) isappropriately prolonged during the whole on-off period. Therefore, thesmaller the input voltage V_(IN) is, the longer the turn-on time T_(ON)of the master switch is consequently. In other words, although the inputvoltage V_(IN) is reduced to increase the on-off frequency value f, theincreasing of the on-off frequency value f is inhibited if the on-timeT_(ON) is prolonged. Therefore, the embodiment of the invention cangreatly ensure the relative steady state of the on-off frequency valuef.

For example, the on-off frequency value f is equal to(2*I_(O)*L*V_(O))/{(V_(IN))2*(T_(ON))²} in a non-continuous DCM mode,where L is the equivalent inductance of the transformer T, and accordingto the scheme provided by the invention, no matter the input voltageV_(IN) is reduced or increased, the value of (V_(IN))²*(T_(ON))² in thefunction is not greatly changed, and the change of the on-off frequencyvalue f can be inhibited, so that the transformer T can be preventedfrom being damaged after being saturated.

Compared with FIG. 6A, the components of the coupling element 106 of thecircuit diagram of FIG. 7A are different than that of FIG. 6A. Thecoupling element 106 is a pulse transformer PT. The circuit and the modefor generating the control signal SQ for the second controller 105 aresimilar to those of FIG. 6A. In this embodiment, the pulse transformerPT is used as a transmission media for data signal interaction betweenthe first controller 104 and the second controller 105, and is providedwith a primary winding LP_(T1) and a secondary winding LP_(T2), in whichthe primary winding LP_(T1) is connected with the second controller 105,and the secondary winding LP_(T2) is connected with the first controller104. The first end of the primary winding LP_(T1) is adapted to receivethe control signal SQ generated by the RS trigger 105 a, and the secondend is coupled with the reference ground potential VSS. The first end ofthe secondary winding LP_(T2) is adapted to generate the first pulsesignal S₁ for driving the master switch Q1, and the second end iscoupled with the ground terminal GND. In the embodiment of FIG. 7A, thecontrol signal SQ is inputted at the first end of the primary windingLP_(T1), and the first pulse signal S₁ is outputted from the first endof the secondary winding LP_(T2). The control signal SQ is transmittedto the input end of the buffer A4, passing through a capacitor C₅₂connected between the output end, which is a node N₅, of the buffer A4and the first end of the primary winding LP_(T1), and the second end ofthe primary winding LP_(T1) is connected with a relatively lowpotential, for example a reference ground potential VSS at a node N₇. Acapacitor C₅₁ is connected between the first end of the secondarywinding LP_(T2) and a signal generation node N_(S) for outputting thefirst pulse signal S₁, and the second end of the secondary windingLP_(T2) is connected with the ground terminal GND at a node N₆. Inaddition, the cathode of a diode D₅₁ is optionally connected with thenode N_(S), and the anode is connected with the ground terminal GND atthe node N₆, and a resistor R₅₁ is optionally connected between the nodeN₅ and the node N₆. The working mechanism of the pulse transformer PT isthat the capacitor C₅₂ is adapted to isolate the direct current, andwhen the control signal SQ is converted to the high level to charge thecapacitor C₅₂, the potential at the first end of the primary windingL_(PT1) is also increased. FIG. 7B shows the waveform of the voltageV_(TX1) of the transmitting interface TX1+ at the first end of theprimary winding L_(PT1), while the node at the second end of the primarywinding L_(PT1) is taken as a transmitting interface TX2−. The controlsignal SQ is transmitted to the secondary winding L_(PT2) through thepulse transformer PT, which increases the potential at the first end ofthe secondary winding L_(PT2). A waveform of the voltage V_(RX1) of thereceiving interface RX1 at the first end of the secondary windingL_(PT2) is as shown in FIG. 7B, and the second end of the secondarywinding L_(PT2) is taken as a receiving interface RX2−. In the process,as the potential of the node N_(S) is also synchronously increased dueto the coupling function of the capacitor C₅₁, the potential of the nodeN_(S) is rapidly increased by the clamping effect of the Schottky diodeD₅₁, and a first pulse signal S₁ at the high level is outputted from thenode N_(S). On the contrary, once the control signal SQ is converted tothe low level, the capacitor C₅₂ discharges through the primary windingL_(PT1), and the capacitor C₅₁ also discharges through the secondarywinding L_(PT2) and the resistor R₅₁, so that the potential of thesignal generation node N_(S) is rapidly dropped, thus the first pulsesignal S₁ at the low level is generated at the signal generation nodeN_(S) and is synchronously changed along with logic state of the controlsignal SQ. The waveform of a second pulse signal S₂, which is an inversephase signal of the first pulse signal S₁, is also shown in FIG. 7B.

The embodiment shown in FIG. 7C is slightly different from that of FIG.7A. As shown in FIG. 7C, one of the feedback voltage V_(FB) and thesensing voltage V_(CS) is inputted into the inverting input terminal ofthe first comparator A1 in the second controller 105; however, in thisembodiment, the feedback voltage V_(FB) is firstly transmitted through afilter 105 g and the sensing voltage V_(CS) is firstly transmittedthrough an amplifier 105 h, then the output of the filter 105 g and theoutput of the amplifier 105 h are combined through an adder 105 i andare further transmitted into the inverting input terminal of the firstcomparator A1. The waveform of an actual ripple voltage at an outputnode N₂₀ as shown in FIG. 1 or FIG. 8, which is described in detaillater, comprises alternating current and direct current, where theaverage voltage of the ripple voltage is equivalent to the voltage ofthe direct current, and the voltage obtained by subtracting the voltageof the direct current from the total ripple voltage is actually equal tothe voltage of the alternating current. The feedback voltage V_(FB) issubstantially a partial voltage of the actual ripple voltage captured atthe output node N₂₀. In addition, the sensing voltage V_(CS) representsthe loading current I_(O), and the direct current of the load currentI_(O), in the alternative-direct current, is much greater than thealternating current of the load current I_(O), so that the averagevoltage of the sensing voltage V_(CS), which also represents analternative-direct current, is equal to the voltage of the directcurrent of the sensing voltage V_(CS). As shown in FIG. 7C, the actualripple voltage is transmitted to a filter 105 g for filtering off thedirect current of the actual ripple voltage and outputting thealternating current. In other words, the voltage of the direct currentof the feedback voltage V_(FB) is subtracted from the total voltage ofthe feedback voltage V_(FB) through the filter 105 g so as the feedbackvoltage V_(FB) only includes the voltage of the alternating current. Inaddition, as shown in FIG. 7C, the voltage drop of the loading currentI_(O), which is the sensing voltage V_(CS) generated by the sensingresistor R_(C), is transmitted to the amplifier 105 h and is outputtedafter being amplified by the amplifier 105 h. The signal outputted fromthe filter 105 g, which is the signal of the alternating currentobtained after the direct current of the feedback voltage V_(FB) isfiltered off by the filter 105 g, and the signal outputted from theamplifier 105 h, which includes both the alternating current and thedirect current and is obtained after the sensing voltage V_(CS) isamplified through the amplifier 105 h, are combined through the adder105 i and are subsequently transmitted into the inverting input terminalof the first comparator A1. Excepting that the feedback voltage V_(FB)or the sensing voltage V_(CS) is not sent directly to the invertinginput terminal of the first comparator A1, the embodiment as shown inFIG. 7C is mostly identical to that as shown in FIG. 7A. In addition,the new feature including the signal outputted from the filter 105 g andthe signal outputted from the amplifier 105 h then combined through theadder 105 i and subsequently inputted into the inverting input terminalof the first comparator A1 is also applied to the embodiments of FIG. 6Aand FIG. 6C.

Referring to FIG. 1 and FIG. 8, the only difference is that the firstend of the secondary winding L_(S) is connected with the output node N₂₀through the rectifier diode D_(REC) and the synchronous switch Q2 inFIG. 1 is omitted in FIG. 8 so that the second end of the secondarywinding L_(S) is directly coupled to the reference ground potential VSS.The anode of the rectifier diode D_(REC) is connected with the first endof the secondary winding L_(S), the cathode is connected with the outputend N₂₀, and the starting voltage ST can be captured at the cathode ofthe rectifier diode D_(REC). Since the synchronous switch Q2 is omitted,the second pulse signal S₂ is not generated. The operation mechanism ofFIG. 8 is similar to that of FIG. 1.

In the voltage converter, if the load 18 is light or empty, the loadcurrent I_(O) is reduced, the on-off frequency value f of the masterswitch Q1 is also reduced correspondingly to the load 18. In addition,the reduction of the on-off frequency value f can be recognized when thevoltage converter makes a sound, for example, if the on-off frequencyvalue f is too low causing the parasitic oscillation, and the noise madefrom a transformer may indicate that the on-off frequency value f isreduced to be about 20 Hz.

FIG. 9 illustrates a circuit diagram of a voltage converter that solvesthe problem of the noise generated by the reduction of the on-offfrequency value f as mentioned above. Referring to FIG. 6A, FIG. 7A orFIG. 7C respectively, the detection signal DE, either the feedbackvoltage V_(FB), the sensing voltage V_(CS), or a detection signal DEoutputted from the adder 105 i, can be adapted to represent thereal-time intensity of the output voltage V_(O) and/or the load currentI_(O) provided to the load 18 and is inputted into the inverting inputterminal of the first comparator A1. In one example referring to FIG.7C, the detection signal DE is inputted into the inverting inputterminal of the first comparator A1, and the first reference voltageV_(REF) is inputted into the non-inverting input terminal of the firstcomparator A1. When the detection signal DE is lower than the firstreference voltage V_(REF), the setting end S of the RS trigger 105 a isset up due to the high level signal outputted from the first comparatorA1, thus the RS trigger 105 a outputs the control signal SQ at the highlevel, and when the high level signal S_(ON) generated by the on-timegenerator 105 c is transmitted to the reset end R of the RS trigger 105a, the RS trigger 105 a outputs the control signal SQ at the low level,which is already specifically described above.

FIG. 9 only shows a portion of the voltage converter, and specificallyshowing the components of the on-time generator 105 c. As shown in FIG.9 and FIG. 10, once the detection signal DE is lower than the firstreference voltage V_(REF), the one-shot trigger 105 b generates thetemporary state pulse signal CLK at the rising-edge of the controlsignal SQ when it jumps from the low level to the high level. FIG. 10illustrates the waveforms taking at two adjacent periods in which thedetection signal DE is lower than the first reference voltage V_(REF)For example, if the detection signal DE, which is the detection signalDE1 in FIG. 10, is lower than the first reference voltage V_(REF) in afirst period TIME1, the voltage converter generates the control signalSQ1 to turn on the master switch Q1 to increase the output voltage V_(O)and/or the load current I_(O), then the detection signal DE is changedto be greater than the first reference voltage V_(REF) at the endingpoint of the first period TIME1, and when the detection signal DE, whichis the detection signal DE2 in FIG. 10, is lower than the firstreference voltage V_(REF) again in a second period TIME2, the voltageconverter generates the control signal SQ2 to turn on the master switchQ1 to increase the output voltage V_(O) and/or the load current I_(O)again. Finally, the detection signal DE is adjusted to be greater thanthe first reference voltage V_(REF) at the ending point of the secondperiod TIME2, and thus the whole cycle is repeated.

As shown in FIG. 10, the detection signal DE1 in the first period TIME1is lower than the first reference voltage V_(REF). At the startingmoment of the first period TIME1, the RS trigger 105 a is set accordingto the high level signal outputted from of the first comparator A1generating the control signal SQ1 at the high level, and at the moment,the control signal SQ1 is converted from the low level to the highlevel, then the one-shot trigger 105 b generates a narrow pulse at thehigh level, or the temporary state pulse signal CKL1, and the process issimilar to that described above with the combination of FIGS. 6A and 7A.The temporary state pulse signal CKL1 generated by the one-shot trigger105 b triggers the on-time generator 105 c to time the on-time T_(ON1),and during the on-time T_(ON1) the master switch Q1 is turned on, thesignal S_(ON1) generated by the third comparator A3 is continuously atthe low level. After the on-time T_(ON1) is ended, the signal S_(ON1)generated by the third comparator A3 is turned to the high level thusresetting the RS trigger 105 a turning the control signal SQ1 to the lowlevel state. As shown in FIG. 10, which only illustrates two on-offperiods of the master switch Q1 for an example, one preset timeT_(SET-)A is started from the starting point of the first period TIME1,after one or multiple on-off periods when the preset time T_(SET-)A isended, the detection voltage DE is greater than the first referencevoltage V_(REF), and the control signal SQ1 is at the low level. Inaddition, the temporary state pulse signal CKL1 is not at the highlevel, thus the capacitor C_(T) has no transient discharge, and thesignal S_(ON1) outputted from the third comparator A3 is kept being atthe high level.

As shown in FIG. 10, after the first period TIME1 is ended, due to thevoltage modulation effect of the voltage converter, the detection signalDE2 is increased to be greater than the first reference voltage V_(REF),and thus the output signal from the first comparator A1 is at the lowlevel. After a time interval, the RS trigger 105 a generates a controlsignal SQ2 at the high level according to the high level output signalof the first comparator A1 at the starting moment of the second periodTIME2 when the detection signal DE2 in the second period TIME2 is lowerthan the first reference voltage V_(REF) again. At this moment, thecontrol signal SQ2 is turned from the low level to the high level, sothat the one-shot trigger 105 b generates a narrow temporary state pulsesignal CKL2 at the high level adapted to trigger the capacitor C_(T) todischarge to a voltage lower than a third reference voltage V_(P), thusthe on-time generator 105 c starts to time the turn-on time T_(ON2), andthe signal S_(ON2) generated by the third comparator A3 is continuouslyat the low level and the master switch Q1 is turned on during theturn-on time T_(ON2). After the turn-on time T_(ON2) is ended, thecapacitor C_(T) is charged to a voltage greater than the third referencevoltage V_(P), and the signal S_(ON2) at the high level generated by thethird comparator A3 in the on-time generator 105 c resets the RS trigger105 a, thus the control signal SQ2 is converted into the low levelstate. During the second period TIME2, as shown in FIG. 10, one presettime T_(SET-B) is started from the starting point of the second periodTIME2, after one or multiple on-off periods when the preset timeT_(SET-B) is ended, the detection voltage DE is greater than the firstreference voltage V_(REF) to meet the load requirements. At this moment,the control signal SQ2 is at the low level, but the temporary statepulse signal CLK2 is not at the high level yet, thus the capacitor C_(T)has no transient discharge, and the signal S_(ON2) outputted from thethird comparator A3 is still at the high level.

As shown in FIG. 9, the output signal, which is either the feedbackvoltage V_(FB), the sensing voltage V_(CS) or the output voltage fromthe adder 105 i, is lower than the first reference voltage V_(REF) inthe period of the preset time T_(SET-)A and present time T_(SET-)B sothat the transformer T can be prevented from making a noise when theon-off frequency value f is too low. As mention above, either thefeedback voltage V_(FB), the sensing voltage V_(CS) or output voltage ofthe adder 105 i is the detection signal DE. Referring to FIG. 9 and FIG.10, the temporary state pulse signal CLK1 is generated when the controlsignal SQ1 during the preset time T_(SET-)A has a frequency value F, andwhen the temporary state pulse signal CLK1 is at the high level withnarrow pulse for more than one times possibly, one or more frequencyvalues F is generated. As shown in FIG. 9, a time generator 113comprises an oscillator 113 a and a frequency divider 113 b, in whichthe oscillator 113 a is adapted to generate an oscillation signaloutputting to the frequency divider 113 b, and the frequency divider 113b is adapted to change the frequency value of the oscillation signal toprovide an upper frequency critical value F_(H) and a lower frequencycritical value F_(L) outputted to a frequency comparator 114 asreference frequency values for comparing with the frequency F of thetemporary state pulse signal CLK1 triggered by the rising-edge of thecontrol signal SQ1. A counter 115 is provided with an additioncalculator and a subtraction counter, and the initial count value of thecounter 115 can be set up in advance. The counter 115 is limited tosubtract 1 from the set initial count value when one frequency value Fis greater than the upper frequency critical value F_(H). The additionor the subtraction is implemented according to the comparison result ofthe frequency comparator 114 transmitted to the counter 115, andcalculation rules defined in advance are executed through the counter115 according to the result. During the preset time T_(SET-)A, dependingon the comparison result of the frequency value F corresponding to thenarrow temporary state pulse signal CLK1 at the high level and areference frequency value, either the counter 115 will add 1 or subtractby 1, and the counter 115 counts for identical times (for example 5times) according to the number of frequency values F (for five differentfrequency values), and finally a total count value can be generated bythe counter 115. In addition, the counter 115 follows some countingconditions, which is an upper critical count value and a lower criticalcount value are defined for the counter 115, once the total count valueexceeds the upper critical count value, it is adjusted to be equal tothe upper critical count value, or when the total count value is lowerthan the lower critical count value, it is adjusted to be equal to thelower critical count value, but when the total count value is equal toone of the upper critical count value and the lower critical countvalue, the total count value is not changed.

In one example, for illustration but not restriction to the embodimentsof the invention, a plurality of narrow temporary state pulse signalsCLK1 at the high level during the preset time T_(SET-)A have fivedifferent frequency values correspondingly, or the total number of thefrequency values F of the temporary state pulse signals CLK1 is five. Inthis situation, the initial count value of the counter 115, which is thelower critical count value, is defined as the binary code elementBIT[00] of two bits, and the upper critical count value is defined as abinary code element BIT[11] of two bits. When the total number of thefrequency values F of the temporary state pulse signals CLK1 is five,each frequency value is compared with the upper critical frequency valueF_(H) and the lower critical frequency value F_(L) in sequence throughthe frequency comparator 114, and the comparison result obtainedincludes a first frequency value lower than the lower critical frequencyvalue F_(L), a second frequency value greater than the upper criticalfrequency value F_(H), a third frequency value lower than the lowercritical frequency value F_(L), a fourth frequency value greater thanthe upper critical frequency value F_(H) and a fifth frequency valuelower than the lower critical frequency value F_(L). As mentioned above,the narrow temporary state pulse signals CLK1 at the high level arecounted by the counter 115, and on the basis of the initial count valueBIT[00], the counter 115 comprises the following counting steps insequence as follows: when the first frequency value is lower than thelower critical frequency value F_(L), the addition counter of thecounter 115 is valid and 1 is added to the comparison result of thefrequency comparator 114; when the second frequency value is greaterthan the upper critical frequency value F_(H), the subtract counter ofthe counter 115 is valid and 1 is subtracted from the comparison resultof the frequency comparator 114; when the third frequency value is lowerthan the lower critical frequency value F_(L), the addition counter ofthe counter 115 is valid and 1 is added to the comparison result of thefrequency comparator 114; when the fourth frequency value is greaterthan the upper critical frequency value F_(H), the subtract counter ofthe counter 115 is valid and 1 is subtracted from the comparison resultof the frequency comparator 114; and when the fifth frequency value islower than the lower critical frequency value F_(L), the additioncounter of the counter 115 is valid and 1 is added to the comparisonresult of the frequency comparator 114. As a result, 1 is added to theinitial count value BIT[00] three times and is subtracted for two times,thereby obtaining the total count value BIT[01]. In another embodiment,when the initial count value BIT[00], the lower critical count valueBIT[00] and the upper critical value BIT[11] mentioned above are notchanged, but the ranges of the five frequency values are changed, on thebasis of the initial count value BIT[00], the counter 115 comprises thefollowing counting steps implemented in sequence as follows: when thefirst frequency value is greater than the upper critical frequency valueF_(H), the subtraction counter of the counter 115 is valid and 1 issubtracted from the comparison result of the frequency comparator 114;when the second frequency value is greater than the upper criticalfrequency value F_(H), the subtraction counter of the counter 115 isvalid and 1 is subtracted from the comparison result of the frequencycomparator 114; when the third frequency value is greater than the uppercritical frequency value F_(H), the subtraction counter of the counter115 is valid and 1 is subtracted from the comparison result of thefrequency comparator 114; when the fourth frequency value is greaterthan the upper critical frequency value F_(H), the subtraction counterof the counter 115 is valid and 1 is subtracted from the comparisonresult of the frequency comparator 114; and when the fifth frequencyvalue is greater than the upper critical frequency value F_(H), thesubtraction counter of the counter 115 is valid and 1 is subtracted fromthe comparison result of the frequency comparator 114. As a result, thetotal count value is less than the lower critical count value BIT[00],so that the final total count value is set of lower critical count valueBIT[00]. In another contrary embodiment, as the initial count valueBIT[00], the lower critical count value BIT[00] and the upper criticalvalue BIT[11] mentioned above are not changed, but the ranges of thefive frequency values are changed, on the basis of the initial countvalue BIT[00], the counter 115 comprises the following counting stepsimplemented in sequence as follows: when the first frequency value islower than the lower critical frequency value F_(L), the additioncounter of the counter 115 is valid and 1 is added to the comparisonresult of the frequency comparator 114; when the second frequency valueis lower than the lower critical frequency value F_(L), the additioncounter of the counter 115 is valid and 1 is added to the comparisonresult of the frequency comparator 114; when the third frequency valueis lower than the lower critical frequency value F_(L), the additioncounter of the counter 115 is valid and 1 is added to the comparisonresult of the frequency comparator 114; when the fourth frequency valueis lower than the lower critical frequency value F_(L), the additioncounter of the counter 115 is valid and 1 is added to the comparisonresult of the frequency comparator 114; and when the fifth frequencyvalue is lower than the lower critical frequency value F_(L), theaddition counter of the counter 115 is valid and 1 is added to thecomparison result of the frequency comparator 114. As a result, thetotal count value is greater than the upper critical count valueBIT[11], so that the final total count value set as the upper criticalcount value BIT[11].

As shown in FIG. 9 and FIG. 10, the frequency values F of the temporarystate pulse signal CLK1 is implemented during the preset time T_(SET-)A,and the total count value from the counter 115 is finally transmittedand encoded/burned into a register 116 for storage. The on-time T_(ON2)during the preset time T_(SET-)B is adjusted relative to the on-timeT_(ON1) during the preset time T_(SET-)A, and the final total countvalue corresponding to the counting frequency value F is used as thebasis for the adjustment of the on-time T_(ON2). The adjustment of theon-time T_(ON2) is illustrated in FIG. 9. As shown in FIG. 9, theon-time generator 105 c mainly comprises a fixed current source 110, twooptional auxiliary current sources 111 and 112, a third switch SW₅₁ anda capacitor C_(T), and the fixed current source 110 and the twoauxiliary current sources 111 and 112 are provided with a workingvoltage through a power supply voltage V_(DD). The current I_(O)outputted from the fixed current source 110 is directly transmitted to anode N_(T) at one end of the capacitor C_(T) to continuously charge thecapacitor C_(T), and the other end of the capacitor C_(T) is connectedwith the ground terminal GND. Furthermore, a fourth switch SW₆₁ isconnected between the auxiliary current source 111 and the node N_(T) atone end of the capacitor C_(T), where the current I₁ outputted from theauxiliary current source 111 is received through one end of the fourthswitch SW₆₁, while the second end of the fourth switch SW₆₁ is connectedwith the node N_(T). When the control end of the fourth switch SW₆₁receives the high level signal, it is turned on, thus the capacitorC_(T) can be charged through the current I₁ outputted from the auxiliarycurrent source 111 at the node N_(T). Similarly, a fifth switch SW₆₂ isconnected between the other auxiliary current source 112 and node N_(T)at one end of the capacitor C_(T), and the current I₂ outputted from theauxiliary current source 112 is received at the first end of the fifthswitch SW₆₂, while the second end is connected with the node N_(T). Whenthe control end of the fifth switch SW₆₂ receives the high level signal,it is turned on, thus the capacitor C_(T) can be charged through thecurrent I2 outputted from the auxiliary current source 112 at the nodeN_(T). The first end of the third switch SW₅₁ is connected with the nodeN_(T), and the second end is connected with the ground terminal GND,thus the third switch SW₅₁ is connected with the capacitor C_(T) inparallel. The temporary state pulse signal CLK1 at high level generatedat the rising-edge of the control signal SQ1 during the preset timeT_(SET-)A in the one-shot trigger 105 b is inputted into the control endof the third switch SW₅₁, thus the third switch SW₅₁ is turned on, andthe capacitor C_(T) is discharged at the node N_(T) when the thirdswitch SW₅₁ is turned on, so that the signal S_(ON1) at the low level isgenerated by the output end of the third comparator A3. After therising-edge of the control signal SQ1, the temporary state pulse signalsCLK1 at the high level with narrow pulse turns back to the low level,and the fixed current source 110 starts to charge the capacitor C_(T) atnode N_(T). Alternatively, if the fourth switch SW₆₁ is turned on, theauxiliary current source 111 and the fixed current source 110 togethercharge the capacitor C_(T) at node N_(T), and if the fifth switch SW₆₂is turned on, the auxiliary current source 112 and the fixed currentsource 110 together charge the capacitor C_(T). The on-time generator105 c is triggered by the temporary state pulse signal CLK1 generated bythe one-shot trigger 105 b to time the on-time T_(ON1), and the signalS_(ON1) generated by the third comparator A3 during the on-time T_(ON1)when the master switch Q1 is turned on is continuously at the low level.While the capacitor C_(T) is charged during the on-time T_(ON1), thevoltage at the node N_(T) of the capacitor C_(T) is greater than thethird reference voltage V_(P), and after the on-time T_(ON1) is ended,the signal S_(ON1) outputted from the third comparator A3 is convertedto the high level during the off-time T_(OFF1), and then the signalS_(ON1) is inputted into the reset end R of the RS trigger 105 a restingthe RS trigger 105 a. The control signal SQ1 generated at the output endQ can drop from the high level to the low level during the off-timeT_(OFF1), and then the master switch Q1 is turned off. If the detectionvoltage DE is still lower than the first reference voltage V_(REF) afterthe first on-off period of the master switch Q1, a second on-off periodis implemented for the master switch Q1, and the operation is repeateduntil the detection voltage DE is greater than the first referencevoltage V_(REF) when the preset time T_(SET-)A is ended. In such anon-off mode, the operation that the master switch Q1 is turned on duringthe on-time T_(ON1) and is turned off during the off-time T_(OFF1) isrepeated for multiple times during the whole preset time T_(SET-)A.

The control signal SQ2 during the preset time T_(SET-)B and the signalCLK2 at the high level with narrow pulse at the rising-edge of thecontrol signal SQ2 are generated from the second controller 105 based onthe total count value of the counter 115 during the preset timeT_(SET-)A. When the on-off frequency value f during the preset timeT_(SET-)A is too low and the transformer T makes the sound, the finaltotal count value of the counter 115 is greater than the preset initialcount value, which is stored in the register 116. The binary codeelement written by the register 116 controls the fourth switch SW₆₁ andthe fifth switch SW₆₂ turning on or off, and when the on-off frequencyvalue f is too low and the total count value is greater than the initialcount value, for example, the total count value is bit BIT[01], orBIT[11], then the total count value is greater than the code elementBIT[00] of the initial count value.

As mentioned above, the total count value BIT[01] is used as the controlsignal of the fourth switch SW₆₁ and the fifth switch SW₆₂, where theon/off state of the fourth switch SW₆₁ is turned on through 0 ofrelatively high bit, and the fifth switch SW₆₂ is turned on through 1 ofrelatively low bit. Furthermore, the total count value BIT[11] is usedas the control signal of the fourth switch SW₆₁ and the fifth switchSW₆₂, in which the fourth switch SW₆₁ is turned on through 1 ofrelatively high bit, and the fifth switch SW₆₂ is turned on through 1 ofrelatively low bit. A schematic diagram of the on-time generator 105 cis illustrated in FIG. 9 as an example, however, other contentwell-known in the art with the control signal data of the registerdecoded by a decoder in advance to subsequently turn on or turn offcorresponding switches through a group of decoding signals can also beimplemented.

When the detection voltage DE is lower than the first reference voltageV_(REF) during the preset time T_(SET-)B, and when the third switch SW₅₁is turned on as the temporary state pulse signal CLK2 is at the highlevel with narrow pulse triggered by the rising-edge of the controlsignal SQ2 during the preset time T_(SET-)B, the capacitor C_(T) isdischarged at node N_(T) through the third switch SW₅₁, so that thesignal S_(ON2) at the low level is generated at the output end of thethird comparator A3. After the rising-edge of the control signal SQ2,the temporary state pulse signals CLK2 at the high level with narrowpulse drops back to the low level, and the fixed current source 110starts to charge the capacitor C_(T at) node N_(T). Alternatively, ifthe fourth switch SW₆₁ is turned on, the auxiliary current source 111and the fixed current source 110 together charge the capacitor C_(T),and if the fifth switch SW₆₂ is turned on, the auxiliary current source112 and the fixed current source 110 together charge the capacitorC_(T). The fourth switch SW₆₁ is controlled to be turned off, and thefifth switch SW₆₂ is thus turned on by the total count value BIT[01] ofthe register 116, so that the current I₂ outputted from the auxiliarycurrent source 112 and the current I₀ outputted from the fixed currentsource 110 are directly transmitted to the node N_(T) at one end of thecapacitor C_(T) to charge the capacitor C_(T). As a result, the chargespeed is relatively fast with the combination of the current I₀ and I₂comparing with that with the single current I₀, as such the capacitorC_(T) is rapidly fully charged in the preset time T_(SET-)B comparing tothat in the preset time T_(SET-)A. Similarly, the fourth switch SW₆₁ andthe fifth switch SW₆₂ are controlled to be turned on by the total countvalue BIT[11] of the register 116, and the current I₁ outputted from theauxiliary current source 111, the current I₂ outputted from theauxiliary current source 112 and the current I₀ outputted from the fixedcurrent source 110 are directly transmitted to the node N_(T) at one endof the capacitor C_(T) to charge the capacitor C_(T). As a result, thecharge speed is relatively fast with the combination of the current I₀,I₁ and I₂ comparing with that of the single current I₀, so that thecapacitor C_(T) can be rapidly fully charged in the preset timeT_(SET-)B relative to that in the preset time T_(SET-)A. The on-timegenerator 105 c is triggered by the temporary state pulse signal CLK2generated by the one-shot trigger 105 b to time the on-time T_(ON2), andthe signal S_(ON2) generated by the third comparator A3 is continuouslyat the low level during the on-time T_(ON2) when the master switch Q1 isturned on. While the capacitor C_(T) is continuously charged during theturn-on time T_(ON2), the voltage of the capacitor C_(T) starts to begreater than the third reference voltage V_(P). After the turn-on timeT_(ON2) is ended, the signal S_(ON2) is converted to the high levelduring the turn-off time T_(OFF2) and is further inputted into the resetend R to reset the RS trigger 105 a, thus the control signal SQ2generated by the output end Q drops back from the high level to the lowlevel during the turn-off time T_(OFF2), and then the master switch Q1is turned off. If the detection voltage DE of the master switch Q1 isstill lower than the first reference voltage V_(REF) after the firston-off period, a second on-off period is implemented for the masterswitch Q1, and the operation is repeated until the detection voltage DEis greater than the first reference voltage V_(REF) after the presettime T_(SET-)B is ended. In the on-off mode, the operation that themaster switch Q1 is turned on in the on-time T_(ON2) and is turned offin the off-time T_(OFF2) can be repeated for multiple times in the wholepreset time T_(SET-)B.

As mentioned above, the current source 111 and/or current source 112 isnot provided in the preset time T_(SET-)A, but the current source 111and/or current source 112 is provided in the preset time T_(SET-)B. As aresult, the charge speed of the capacitor C_(T) is relatively fastbecause the total current during the on-time T_(ON2) of the preset timeT_(SET-)B is larger, so that it takes shorter time for the voltage atthe node N_(T) being greater than the third reference voltage V_(P), andthus the on-time T_(ON2) is shorter than the on-time T_(ON1).Considering that the on-off frequency value f of the master switch Q1 isreduced as the on-time T_(ON) increases and is increased as the on-timeT_(ON) decreases. As such, when the load 18 is a light load or emptyload, the on-off frequency value f in the on-time T_(ON1) is increasedwhen the on-time T_(ON2) is reduced, and thus the transformer T can beprevented from making a sound.

Actually, the relative amounts of the on-time T_(ON1) and the turn-ontime T_(ON2) are closely associated with the initial count value of thecounter 115. For example, if the initial count value of the counter 115in the preset time T_(SET-)A is BIT[01] or BIT[10], one of the fourthswitch SW₆₁ and the fifth switch SW₆₂ is turned on and the other one isturned off, then the capacitor C_(T) is charged by the current I₁outputted from the auxiliary current source 111 or the current I₂outputted from the auxiliary current source 112 together with thecurrent I₀ of the fixed current source 110 in the on-time T_(ON1), i.e.,the total charge current is (I₁+I₀) or (I₂+I₀. On the basis of theinitial count value, for example BIT[01], the counter 115 operates withthe following counting steps with different frequency values as follows:when the first frequency value is greater than the upper criticalfrequency value F_(H), the subtraction counter of the counter 115 isvalid and 1 is subtracted from the comparison result of the frequencycomparator 114; when the second frequency value is lower than the lowercritical frequency value F_(L), the addition counter of the counter 115is valid and 1 is added to the comparison result of the frequencycomparator 114; when the third frequency value is greater than the uppercritical frequency value F_(H), the subtraction counter of the counter115 is valid and 1 is subtracted from to the comparison result of thefrequency comparator 114; when the fourth frequency value is lower thanthe lower critical frequency value F_(L), the addition counter of thecounter 115 is valid and 1 is added to the comparison result of thefrequency comparator 114; and when the fifth frequency value is greaterthan the upper critical frequency value F_(H), the subtraction counterof the counter 115 is valid and 1 is subtracted from the comparisonresult of the frequency comparator 114. When the final count value isBIT[00] and the total charge current of the capacitor C_(T) is I0 in theturn-on time T_(ON2), the total charge time of the capacitor C_(T) inthe on-time T_(ON2) is greater than that in the on-time T_(ON1),equivalently, the on-time T_(ON2) is adjusted to be greater than theon-time T_(ON1), and thus the on-off frequency value f can be adjustedto a small value in the preset time T_(SET-)B from a large value in thepreset time T_(SET-)A.

In the summary, the control signal SQ1 of the second controller 105 ofthe secondary winding is transmitted to the first controller 104 of theprimary winding through the coupling element 106 in the preset timeT_(SET-)A as shown in FIG. 10, so that the first pulse signal S₁generated by the first controller 104 is enabled to control the masterswitch Q1 turning on during on-time T_(ON1) in the on-off period. Asshown in FIG. 10, the control signal SQ2 of the second controller 105 ofthe secondary winding is transmitted to the first controller 104 of theprimary winding through the coupling element 106 in the preset timeT_(SET-)B, so that the first pulse signal S₁ generated by the firstcontroller 104 is enabled to control the master switch Q1 turning onduring on-time T_(ON2) in the on-off period. When the final total countvalue obtained by calculating the number of the frequency values F ofthe CLK1 triggered by the rising-edge of the control signal SQ1 by thecounter 115 in the preset time T_(SET-)A is greater than the initialcount value, the on-time Torn during the preset time T_(SET)-B is lessthan the on-time T_(ON1). Vice versa, when the final total count valueis less than the initial count value, the on-time T_(ON2) during thepreset time T_(SET-)B is greater than the on-time T_(ON1). When thefinal total count value is equal to the initial count value, the on-timeT_(ON2) during the preset time T_(SET-)B is equal to the on-timeT_(ON1). The reason is that when the detection voltage DE is lower thanthe first reference voltage V_(REF), the total count value can beupdated once, and whether the switches SW₆₁ and SW₆₂ are turned on ornot is directly determined by the code element in the total count value,therefore, when the detection voltage DE is lower than the firstreference voltage V_(REF) in a latter time, the on-time is determined bythe total count value of the previous time. In the present invention,the code elements only includes two bits and the two extra auxiliarycurrent sources 111 and 112 are provided for example, in practicaltopology, the initial count value, the upper critical count value andthe lower critical count value are not limited by only two bits codeelements of two bits, and the number of the auxiliary current sources isnot limited by only two currents.

The above embodiments describe the structure and operation mechanism ofthe voltage converters using the first pulse signal S₁ driving themaster switch Q1 to switch on/off and the second pulse signal S₂ drivingthe synchronous switch Q2 to switch on/off.

In the present invention, a data transmission medium between the firstand second controllers 104 and 105, i.e., the coupling element 106, isvery important. In one example, the coupling element 106 includes apulse transformer PT, and the structure of the pulse transformer PT isdescribed in FIG. 11A to FIG. 13C.

As shown in FIG. 11A, a pulse transformer PT of the voltage converter,which is only a portion of a whole PCB includes a circuit board 200 withall the electronic devices surface-mounted to the circuit board. A firstthrough hole 201 and a second through hole 202 penetrating through thethickness of the circuit board 200 are formed side by side on thecircuit board 200 by drilling, etching or laser cutting and the likes. Astrip-shaped gap 203 penetrating through thickness of the circuit board200 is optionally formed in the region of the circuit board 200 betweenthe first through hole 201 and the second through hole 202. Optionally,the first through hole 201 and the second through hole 202 aresymmetrically disposed at the two opposite sides of the gap 203 bytaking the gap 203 as the central symmetric line, and the first throughhole 201 and the second through hole 202 can be squares. A helical coil202 a is formed around the first through hole 201 on the surface of thecircuit board 200 serving as a primary winding of the pulse transformerPT. The helical coil 202 a includes multiple concentric squareconducting rings surrounding the first through hole 201, and each of theconducting rings are placed on the same plane of the circuit board 200.The central position of the helical coil 202 a and the central positionof the first through hole 201 are approximately overlapped. Similarly,another helical coil 202 b is formed around the second through hole 202on the same surface of the circuit board 200 serving as a secondarywinding of the pulse transformer PT. The helical coil 202 b includesmultiple concentric square conducting rings surrounding the secondthrough hole 202, and each of the conducting rings are placed on thesame plane of the circuit board 200. The central position of the helicalcoil 202 b and the central position of the second through hole 202 areapproximately overlapped. The helical coil 202 a has a head end and atail end. Similarly, the helical coil 202 b has a head end and a tailend. In one embodiment, the multiple concentric square conducting ringsof the helical coil 202 a can be formed by forming a helical shallowtrench on the upper surface or the lower surface of the circuit board200 surrounding the first through hole 201 including a plurality ofconcentric square grooves from inside to outside filled with conductingmaterials, for example, metal copper or the like. Similarly, themultiple concentric square conducting rings of the helical coil 202 bcan be formed by forming a helical shallow trench on the upper surfaceor the lower surface of the circuit board 200 surrounding the secondthrough hole 202 filled with conducting materials. In other embodiments,the helical coil 202 a and 202 b can be formed by directly mounting aseries of multiple concentric square metal coils to the upper surface ofthe circuit board 200 by adhering, depositing, sputtering,electroplating and the like; for example, they are made by plating metalwiring or wire TRACE on the circuit board 200. In one embodiment shownin FIG. 11A, the helical coil 202 a or 202 b is a square. However, thecoils of the helical coil 202 a or 202 b may also be a series ofconcentric rings or various polygon shapes and the like (not shown). Asshown FIG. 11A, the helical coil 202 a or 202 b includes only a singlelayer, however, in other embodiments, the helical coil 202 a includesstacked multilayer helical coils such that the helical coils atdifferent layers are disposed in separated planes parallel with eachother surrounding the first through hole 201 (not shown). Similarly, thehelical coil 202 b can include stacked multilayer helical coils, suchthat the helical coils at different layers are disposed in separatedplanes parallel with each other surrounding second through hole 202 (notshown). In the multilayer helical coil structure, the helical coils atdifferent layers are electrically isolated by an insulating layerlaminated between two helical coil layers but any two adjacent helicalcoils need to interconnect as follows: the second end (or ending end) ofone helical coil and the first end (or beginning end) of the followingadjacent helical coil need to be electrically connected through aninterconnection wire connecting these multilayer helical coils inseries. For example, in the multilayer helical coil, the first end (orthe beginning end) of the first helical coil in the topmost layer isserved as one terminal of the series structure of a plurality of helicalcoils, and the ending end of the last helical coil in the bottommostlayer is served as the other terminal of the series structure of theplurality of helical coils.

As shown in FIG. 11A, the pulse transformer PT includes a U-shapedmagnetic core 210 and a stripe-shaped magnetic core 211. The magneticcore 210 includes two side portions 210 a and portion 210 b extending intwo parallel planes and a middle portion 210 c perpendicular to the sideportions 210 a and 210 b with each of the side portions 210 a and 210 brespectively connected to each end side of the middle portion 210 c.Substantively, both side portions 210 a and 210 b and the middle portion210 c are integrated in one piece forming U-shaped magnetic core 210.The side portion 210 a of the U-shaped magnetic core 210 is insertedinto the first through hole 201 while the side portion 210 b of theU-shaped magnetic core skeleton 210 is accordingly inserted into thesecond through hole 202, so that the magnetic core 210 is mounted on thecircuit board 200. Furthermore, in order to form a closed magneticcircuit loop, the magnetic core 211 also needs to be attached with themagnetic core 210. In FIG. 11B, the magnetic core 210 is inserted fromthe front side of the circuit board 200, while the respective front endfaces of the two side portions 210 a and 210 b of the magnetic core 210are tightly attached to one surface of the magnetic core 211 on theother side of the circuit board 200, thus building the magnetic circuit.A gap 204 is reserved between the side face of one side portion 210 a ofthe magnetic core 210 and the side wall of the first through hole 201and between the side face of the side portion 210 b of the magnetic core210 and the side wall of the second through hole 202. In FIG. 11B, sincethe magnetic core 210 and the magnetic core 211 are attached together.It is possibly to break the magnetic cores 210 and 211 from the circuitboard 200 if the electronic device with the pulse transformer PTbuilt-in is shook or falls off. Preferably, some insulating glue isapplied on the circuit board 200 to glue or firmly hold the magneticcores 210 and 211 on the circuit board 200 without shifting. The printedcircuit board 200 is used to mount the transformer T, a chip packageintegrated with the first controller 104 and a chip package integratedwith the second controller 105 or the like, as such certain regions forthose devices on the circuit board 200 are reserved before forming thefirst through hole 201 and the second through hole 202. The masterswitch Q1 and the synchronous switch Q2 may be externally mounted on thePCB circuit board 200, or the master switch Q1 and the first controller104 may be integrated in one chip package and then mounted on the PCBcircuit board 200, and/or the synchronous switch Q2 and the secondcontroller 105 are integrated in one chip package and then mounted onthe PCB circuit board 200.

FIG. 12A shows another structure of the pulse transformer PT, which alsoincludes the U-shaped magnetic core 210, a rectangular or squaremagnetic core 211, a first chip package 301 and a second chip package302 instead of the helical coils 202 a and 202 b shown in FIG. 11A. Theflat square first chip package 301 includes a first central hole 314penetrating through the thickness of the first chip package 301 on aposition relatively close to the central position and at least two pins312 and 313 configured for butt-welding with a pad on the circuit board200, for example, by the tin welding surface placement technology. Theflat square second chip package 302 includes a second central hole 324penetrating through the thickness of the second chip package 302 on aposition relatively close to the central position and at least two pins322 and 323 configured for butt-welding with a pad on the circuit board200. In this embodiment, the adjacent first through hole 201 and secondthrough hole 202 are also formed on the circuit board 200. When thefirst chip package 301 and second chip package 302 are mounted on thecircuit board 200, the first central hole 314 and the second centralhole 324 are aligned with the first through hole 201 and second throughhole 202 of the circuit board 200 respectively. The first central hole314 and the second central hole 314 are overlapped with the firstthrough hole 201 and the second through hole respectively, as such theside portion 210 a of the U-shaped magnetic core 210 is easily to insertthrough the first central hole 314 and the first through hole 201 andthe side portion 210 b of the U-shaped magnetic core 210 accordinglyinserts through the second central hole 324 and the second through hole202. In FIG. 12B, the magnetic core 211 and the magnetic core 210 areattached together, where the magnetic core 210 is inserted from thefront side of the circuit board 200, while the respective front endfaces of two side portions 210 a and 210 b of the magnetic core 210 aretightly jointed with one surface of the magnetic core skeleton 211 onthe other side of the circuit board 200, thus building the magneticcircuit. As shown in FIG. 12B, a gap 204 also is reserved between theside face of one side portion 210 a of the magnetic core 210 and therespective side walls of the first through hole 201 and the firstcentral hole 314, and gap 204 is also reserved between the side face ofanother side portion 210 b of the magnetic core 210 and the respectiveside walls of the second through hole 202 and the second central hole324.

In the embodiment of FIG. 12A, both the first chip package 301 and thesecond chip package 302 are independent chips and attached on thecircuit board 200 separately. In the embodiment of FIG. 12C-1, the firstchip package 301 and the second chip package 302 are integrated in onepiece that is mounted on the circuit board 200. In the top view of FIG.12C-2, the first chip package 301 and the second chip package 302 arearranged side by side, where one corner portion 311 a of the first chippackage 301 and one corner portion 321 a of the second chip package 302are close to each other, and the two chips are connected togetherthrough a connecting portion 331. Another corner portion 311 b of thefirst chip package 301 and another corner portion 321 b of the secondchip package 302 are close to each other, and the two chip packages areconnected together through a connecting portion 332. Alternatively, theconnecting portions 331 and 332 may be in other positions between thefirst and second chip packages as long as the interconnected first chippackage 301 and second chip package 302 are substantially coplanar andcan be synchronously mounted on the circuit board 200.

FIG. 12D is a perspective view of the structure shown in FIG. 12A withthe wiring. The first chip package 301 includes a helical wiring 315while the second chip package 302 includes a helical wiring 325, and theshapes of the helical wirings 315 and 325 are shown in FIG. 12E for anexample. In FIG. 12E, one base plate 317 is optionally used to supportone silicon substrate 316, however the substrate 316 can also be usedsolely. Each of the base plate 317 and the substrate 316 include a holeat the respective central position. The helical wiring 315 is formed onthe upper surface of the substrate 316 surrounding the central holes ofthe substrate 316 and/or the base plate 317. Because the helical wiring315 is a conductor, the helical wiring 315 is electrically insulatedfrom the substrate 316 by an insulating layer. Similarly, anothersubstrate 326 arranged side by side with the substrate 316 and isoptionally supported by one base plate 327, but the substrate 326 may beused solely. Each of the base plate 327 and the substrate 326 include ahole at the respective central position. The helical wiring 325 isformed on the upper surface of the substrate 326 surrounding the centralholes of the substrate 326 and/or base plate 327. Because the helicalwiring 325 is a conductor, the helical wiring 325 is electricallyinsulated from the substrate 326 through an insulating layer. The baseplates 317 and 327 can be a metal lead frame and the like. In FIG. 12E,the helical wiring 315 or 325 is only a single-layer, however, in otherembodiments, the helical wiring 315 or 325 can be stacked multilayerhelical wirings formed on the substrate 316 or 326, so that the helicalwirings at different layers are arranged in planes parallel to eachother and surrounding the central hole 314 or 324. The helical coils atdifferent layers in the multilayer helical coils are electricallyinsulated by a dielectric layer (for example, silicon dioxide, and thelike), but any two adjacent helical coils are interconnected as follows:the second end (or the ending end) of one helical coil and the first end(or beginning end) of the following adjacent helical coil areelectrically connected through an interconnection wire, as such thesemultilayer helical coils are connected in series. In addition, the firstend (or the beginning end) of the first helical coil in the topmostlayer is served as one terminal of the series of the plurality ofhelical coils, and the second end (or the ending end) of the lasthelical coil in the bottommost layer is served as another terminal ofthe series of the plurality of helical coils.

As shown in FIG. 12D, the first chip package 301 has a plastic packagebody 311, and the second chip package 302 has a plastic package body321. In the first chip package 301, the plastic package body 311encapsulates the substrate 316 and/or base plate 317 and the helicalwiring 315 formed on the upper surface of the substrate 316 therein. Alead 318, which is formed for example by wire bonding, connects one endof the helical wiring 315 and the pin 312 formed on the substrate 316,and another lead 318 connects the other end of the helical wiring 315and the pin 313 formed on the substrate 316. The leads 318 are alsoencapsulated inside the plastic package body 311. A portion of each ofthe pins 312 and 313 connecting to the lead 318 is coated by the plasticpackage body 311, but another portion of each of the pins 312 and 313 isextending out of the plastic package body 311 for butt-welding with thepads on the circuit board 200. Similarly, in the second chip package302, the plastic package body 321 encapsulates the substrate 326 and/orthe base plate 327 and the helical wiring 325 formed on the uppersurface of the substrate 326. A lead 328, which is also formed by wirebonding, connects one end of the helical wiring 325 and the pin 322formed on the substrate 326, and another lead 328 connects the other endof the helical wiring 325 and the pin 323 formed on the substrate 326.Similarly, the lead 328 is also encapsulated inside the plastic packagebody 321. A portion of each of the pins 322 and 323 connecting to thelead 318 are encapsulated by the plastic package body 311, but anotherportion of each of the pins 322 and 323 is extending out of the plasticpackage body 311 respectively for butt-welding with the pad formed onthe circuit board 200. The plastic package bodies 311 and 321 can bemade of materials like epoxy resin.

As shown in FIG. 12D, in the first chip package 301, the first centralhole 314 penetrates through the thicknesses of the plastic package body311, the substrate 316 and/or the base plate 317 and is substantiallylocated at the central positions of the plastic package body 311, thesubstrate 316 and/or the base plate 317. The helical wiring 315, or theseries of concentric square conducting rings, surround the first centralhole 314 is served as the primary winding of the pulse transformer PT.Similarly, in the second chip package 302, the second central hole 324penetrates through the thicknesses of the plastic package body 321, thesubstrate 326 and/or the base plate 327 and is substantially located atthe central positions of the plastic package body 321, the substrate 326and/or the base plate 327. The helical wiring 325, or the series ofconcentric square conducting rings, surround the second central hole 324is served as the secondary winding of the pulse transformer PT. Withrespect to the embodiments in FIG. 12C-1 and FIG. 12C-2, in the MOLDINGstep, the plastic package body 311 of the first chip package 301 and theplastic package body 321 of the second chip package 302 aresynchronously and integrally molded in one whole piece. One cornerportion 311 a of the plastic package body 311 and one corner portion 321a of the plastic package body 321 are close to each other and areconnected together through a connecting portion 331. Another cornerportion 311 b of the plastic package body 311 and one corner portion 321b of the plastic package body 321 are close to each other and areconnected together through a connecting portion 332. In the embodimentof FIG. 11B, a strip-shaped gap 203 may or may not be prepared inbetween the first through hole 201 and the second through hole 202 onthe circuit board 200. In the embodiments of FIG. 12A to FIG. 12E, themiddle portion 210 c of the magnetic core 210 and the magnetic core 211are parallel to the respective planes of the first chip package 301, thesecond chip package 302, and the circuit board 200, thus the sideportion 210 a and the side portion 210 b of the magnetic core 210 areperpendicular to the respective planes of the first chip package 301,the second chip package 302, and the circuit board 200. When the firstchip package 301 and the second chip package 302 are mounted on thecircuit board 200, the substrate 316 and/or the base plate 317, thesubstrate 326 and/or the base plate 327, as well as the plastic packagebodies 311 and 321 are all parallel with the circuit board 200.

FIG. 13A shows another structure of the pulse transformer PT including afirst chip package 401 having a U-shaped magnetic core 410 and a secondchip package 402 having a U-shaped magnetic core 420. In the first chippackage 401, as shown in FIG. 13B, the magnetic core 410 includes a sideportion 410 a and a side portion 410 c parallel to each other and amiddle portion 410 b perpendicular to and connecting the side portions410 a and 410 c. One first coil 415 is wound around the middle portion410 b and is electrically connected with a pin 412 directly at one witha pin 413 directly at the other end, where the pins 412 and 413 areadjacent to the magnetic core 410. The plastic package body 411encapsulates the magnetic core 410 and the first coil 415, in which aportion of the pin 412 connecting to the first coil winding 415 isencapsulated by the plastic package body 411, but another portion of thepin 412 is extending out of the plastic package body 411 forbutt-welding with a pad on the circuit board 200. Similarly, a portionof the pin 413 connecting to the first coil winding 415 is encapsulatedby the plastic package body 411, but another portion of the pin 413 isextending out of the plastic package body 411 for butt-welding with apad on the circuit board 200. In the second chip package 402, as shownin FIG. 13B, the magnetic core 420 includes a side portion 420 a and aside portion 420 c parallel to each other, and a middle portion 420 bperpendicular to and connecting the side portions 420 a and 420 ctogether. One second coil 425 is wound around the middle portion 420 bhaving one end of the second coil 415 electrically connected with a pin422 directly and the other end of the second coil winding 425electrically connected with a pin 423 directly, where the pins 422 and423 are adjacent to the magnetic core 420. The plastic package body 421encapsulates the magnetic core 420 and the second coil 425. A portion ofthe pin 422 connecting to the second coil winding 425 is encapsulated bythe plastic package body 421, but the other portion of the pin 422 isextending out of the plastic package body 421 for butt-welding with thepad on the circuit board 200. Similarly, a portion of the pin 423connecting to the second coil winding 425 is encapsulated by the plasticpackage body 421, but another portion of the pin 423 is extending out ofthe plastic package body 421 for butt-welding with the pad on thecircuit board 200. In the embodiments of FIG. 13A to FIG. 13C, themiddle portion 410 b and the side portions 410 a and 410 c of themagnetic core 410 are coplanar and parallel to the plane of the firstchip package 401, and the middle portion 420 b and the side portions 420a and 420 c of the magnetic core 420 are coplanar and parallel to theplane of the second chip package 402. Moreover, when the first chippackage 401 and the second chip package 402 are mounted on the circuitboard 200 side by side, the magnetic core 410, the magnetic core 420 andthe corresponding plastic package bodies 411 and 421 are all parallelwith the circuit board 200.

As shown in FIG. 13A, the front end faces 410 a-1 and 410 c-1 of theside portion 410 a and 410 c of the magnetic core 410 are required to beexposed from the side face 411 a of the plastic package body 411. Thefront end faces 410 a-1 and 410-c are actually the cutting faces of theside portions 410 a and 410 c and perpendicular to the length directionof the side portions 410 a and 410 c respectively. Similarly, the frontend faces 420 a-1 and 420 c-1 of the side portions 420 a and 420 c ofthe magnetic core 420 are required to be exposed from the side faces 421a and 421 c of the plastic package body 421. The front end faces 420 a-1and 420 c-1 are actually the cutting faces of the side portion 420 a and420 c and perpendicular to the length direction of the side portions 420a and 420 c. When the pulse transformer PT is used, the side face 411 aof the plastic package body 411 and the side face 421 a of the plasticpackage body 421 are facing to each other to enable the front end faces410 a-1 and 410 c-1 of the side portion 410 a of the magnetic core 410to be aligned and contacted with the front end faces 420 a-1 and 420 c-1of the side portion 420 a of the magnetic core 420 respectively, thusforming a closed magnetic core circuit between the two magnetic cores410 and 420 along the side portion 410 a of the magnetic core 410 to theside portion 420 a of the magnetic core 420 and along the side portion420 c of the magnetic core 420 to the side portion 410 c of the magneticcore 410.

FIG. 13B shows a closed structure of the pulse transformer PT of FIG.13A. When the first chip package 401 and the second chip package 402 aremounted on the circuit board 200 close to each other with the side face411 a of the plastic package body 411 of the first chip package 401 incontact with one side face 421 a of the plastic package body 421 of thesecond chip package 402, as such the front end face 410 a-1 of the sideportion 410 a of the magnetic core 410 and the front end face 420 a-1 ofthe side portion 420 a of the magnetic core 420 are joined together.Similarly, the front end face 410 c-1 of the side portion 410 c of themagnetic core 410 and the front end face 420 c-1 of the side portion 420c of the magnetic core 420 are jointed together. As a result, themagnetic core 410 and the magnetic core 420 are joined together formingan annular magnetic core structure.

The structure of the transformer PT in FIG. 13C is slightly differentwith that in FIG. 13B. In FIG. 13B, the side face 411 a of the plasticpackage body 411 and the side face 421 a of the plastic package body 421are jointed completely. In FIG. 13C, when the first chip package 401 andthe second chip package 402 are mounted on the circuit board 200 side byside close to each other with but a gap 430 formed between the side face411 a of the plastic package body 411 and the side face 421 a of theplastic package body 421. Similar with the structure in FIG. 13B, theside face 411 a of the plastic package body 411 of the first chippackage 401 and the side face 421 a of the plastic package body 421 ofthe second chip package 402 are aligned and facing each other, thus thefront end face 410 a-1 of the side portion 410 a of the magnetic core410 and the front end face 420 a-1 of the side portion 420 a of themagnetic core 420 are aligned and facing each other, and the front endface 410 c-1 of the side portion 410 c of the magnetic core 410 and thefront end face 420 c-1 of the side portion 420 c of the magnetic core420 are aligned and facing each other. With the side portion 410 a ofthe magnetic core 410 and the side portion 420 a of the magnetic core420 aligned with the gap existing there between and the side portion 410c of the magnetic core 410 and the side portion 420 c of the magneticcore 420 aligned with the gap existing there between, the magnetic core410 and the magnetic core 420 are jointed forming an annular magneticcore structure. In this embodiment, the side portions 410 a and 410 c ofthe magnetic core 410 and the side portions 420 a and 420 c of themagnetic core 420 are disconnected with an air gap formed in between forpreventing magnetic saturation. Since the permeability of air is only afew thousandths of the permeability of an iron core for example, theaverage permeability of the magnetic core with the air gap decreasessignificantly, thus all the magnetic flux are nearly dropped with themagnetic core with the air gap. In this case, not only the residualmagnetic flux density will be reduced, but also the maximum magneticflux density may reach a saturation level, so that the magnetic fluxincrement is increased, and magnetic saturation will not occur to themagnetic core of the transformer. In this embodiment, an insulatingmaterial 450 is optionally filled in the gap 430 between the side face411 a of the plastic package body 411 and the side face 421 a of theplastic package body 421. The insulating material 450 not only achievethe electrical isolation, but also can effectively enhance the bondstrength of firmly adhering the first chip package 401 and the secondchip package 402 on the circuit board 200.

The typical embodiments of specific structures of the detaileddescription are provided through the explanation and drawings above, andthe foregoing invention proposes present preferred embodiments, butthese contents are not intended to limit the invention. Various changesand amendments will be apparent for those skilled in the art afterreading the explanation above. Therefore, the appended claims shall bedeemed to cover all changes and amendments of the real intention andscope of the invention. Any equivalent range and content within theclaims shall all fall within the intention and scope of the invention.

1. A pulse transformer comprising: a first U-shaped magnetic corecomprising a first side portion comprising a front end surface; and asecond side portion parallel to the first side portion, the second sideportion comprising a front end surface; a second strip-shaped magneticcore; and a printed circuit board having a first side and a second side,the printed circuit board comprising a first through hole and a secondthrough hole penetrating through a thickness of the printed circuitboard; wherein the first side portion of the first U-shaped magneticcore is inserted into the first through hole of the printed circuitboard from the first side of the printed circuit board to the secondside of the printed circuit board; wherein the second side portion ofthe first U-shaped magnetic core is inserted into the second throughhole of the printed circuit board from the first side of the printedcircuit board to the second side of the printed circuit board; whereinthe front end surface of the first side portion of the first U-shapedmagnetic core is directly attached to a surface of the secondstrip-shaped magnetic core located on the second side of the printedcircuit board; and wherein the front end surface of the second sideportion of the first U-shaped magnetic core is directly attached to thesurface of the second strip-shaped magnetic core.
 2. The pulsetransformer of claim 1, wherein a surface of the first side or a surfaceof the second side of the printed circuit board comprises a firsthelical coil and a second helical coil; wherein the first helical coilis planarized; wherein the second helical coil is planarized; whereinthe first helical coil surrounds the first through hole; and wherein thesecond helical coil surrounds the second through hole.
 3. The pulsetransformer of claim 1, wherein a gap penetrating through the thicknessof the printed circuit board is disposed between the first and secondthrough holes on the printed circuit board; wherein the first and secondthrough holes are symmetric with respect to a centerline of the gap. 4.The pulse transformer of claim 1, wherein a surface of the first side ora surface of the second side of the printed circuit board comprises afirst helical coil and a second helical coil; wherein the first helicalcoil is planarized; wherein the second helical coil is planarized;wherein the first helical coil surrounds the first through hole; whereinthe second helical coil surrounds the second through hole; and whereinthe first and second helical coils are of square shapes or roundedshapes.
 5. The pulse transformer of claim 1, wherein insulating glue iscoated on the printed circuit board for mounting the first U-shapedmagnetic core and the second strip-shaped magnetic core on the printedcircuit board.
 6. The pulse transformer of claim 1, wherein a surface ofthe first side or a surface of the second side of the printed circuitboard comprises a first helical coil and a second helical coil; whereinthe first helical coil is planarized; wherein the second helical coil isplanarized; wherein the first helical coil surrounds the first throughhole; wherein the second helical coil surrounds the second through hole;wherein the first helical coil comprises a first end as a first terminaland a second end as a second terminal; and wherein the first helicalcoil is in series between the first end and the second end.
 7. The pulsetransformer of claim 1, wherein a surface of the first side or a surfaceof the second side of the printed circuit board comprises a firsthelical coil and a second helical coil; wherein the first helical coilis planarized; wherein the second helical coil is planarized; whereinthe first helical coil surrounds the first through hole; wherein thesecond helical coil surrounds the second through hole; wherein thesecond helical coil comprises a first end as a first terminal and asecond end as a second terminal; and wherein the second helical coil isin series between the first end and the second end.
 8. The pulsetransformer of claim 1, wherein the printed circuit board furthercomprises: a power level main transformer, wherein a primary winding ofthe power level main transformer receives an input voltage and providesan output voltage for a load at a secondary winding, and wherein theprimary winding of the power level main transformer is connected with amaster switch in series; a first semiconductor chip comprising a firstcontroller for producing a first pulse signal to drive the master switchto be turned on or turned off; and a second semiconductor chipcomprising a second controller for comparing a detection voltagerepresenting an output voltage value and/or representing a load currentvalue with a first reference voltage, thus determining a logic state ofa control signal produced thereof according to a comparison result;wherein the pulse transformer transfers the logic state of the controlsignal to the first controller, so that the first controller determinesthe logic state of the first pulse signal according to the logic stateof the control signal, thus determining to turn on or turn off themaster switch.
 9. A pulse transformer comprising: a first U-shapedmagnetic core comprising a first side portion comprising a front endsurface; and a second side portion parallel to the first side portion,the second side portion comprising a front end surface; a secondstrip-shaped magnetic core; wherein the pulse transformer is installedon a printed circuit board comprising first and second through holespenetrating through a thickness of the printed circuit board; a firstchip package comprising a first central hole; and a second chip packagecomprising a second central hole; wherein the first and second chippackages are installed on the printed circuit board; wherein the firstcentral hole is aligned to overlap with the first through hole; whereinthe second central hole is aligned to overlap with the second throughhole; wherein the first side portion of the first U-shaped magnetic coreis inserted into the first central hole and the first through hole froma first side of the printed circuit board to a second side of theprinted circuit board; wherein the second side portion of the firstU-shaped magnetic core is inserted into the second central hole and thesecond through hole; wherein the front end surface of the first sideportion of the first U-shaped magnetic core is directly attached to asurface of the second strip-shaped magnetic core located on the secondside of the printed circuit board; and wherein the front end surface ofthe second side portion of the first U-shaped magnetic core is directlyattached to the surface of the second strip-shaped magnetic core. 10.The pulse transformer of claim 9, wherein the first chip packagecomprises: a first substrate having a first helical wiring formed on asurface; two pins disposed near the first substrate, wherein two ends ofthe first helical wiring are correspondingly connected to the two pinsthrough a lead respectively; and a first plastic package bodyencapsulating the first substrate, the first helical wiring and thelead; wherein a portion of each pin of the two pins is covered by thefirst plastic package body, but another portion of each pin of the twopins extends out of the first plastic package body for welding with arespective pad on the printed circuit board; wherein the first centralhole penetrates through the first plastic package body and the firstsubstrate; and wherein the first helical wiring surrounds the firstcentral hole.
 11. The pulse transformer of claim 10, wherein the firsthelical wiring comprises a first end as a first terminal and a secondend as a second terminal; and wherein the first helical wiring is inseries between the first end and the second end.
 12. The pulsetransformer of claim 9, wherein the second chip package comprises: asecond substrate having a second helical wiring formed on a surface; twopins disposed near the second substrate, wherein two ends of the secondhelical wiring are correspondingly connected to the two pins through alead respectively; and a second plastic package body encapsulating thesecond substrate, the second helical wiring and the lead; wherein aportion of each pin of the two pins is covered by the second plasticpackage body, but another portion of each pin of the two pins extendsout of the second plastic package body for welding with a respective padon the printed circuit board; wherein the second central hole penetratesthrough the second plastic package body and the second substrate; andwherein the second helical wiring surrounds the second central hole. 13.The pulse transformer of claim 12, wherein the second helical wiringcomprises a first end as a first terminal and a second end as a secondterminal; and wherein the second helical wiring is in series between thefirst end and the second end.
 14. The pulse transformer of claim 9,wherein the first U-shaped magnetic core and the second strip-shapedmagnetic core are attached to the printed circuit board by insulatingglue.
 15. The pulse transformer of claim 9, wherein the first chippackage and the second chip package are connected through one or moreconnecting portions forming a coplanar integrated structure forinstalling the first chip package and the second chip package on theprinted circuit board.
 16. The pulse transformer of claim 9, wherein theprinted circuit board comprises: a power level main transformercomprising a primary winding and a secondary winding; wherein theprimary winding of the power level main transformer receives an inputvoltage and provides an output voltage for a load at the secondarywinding; wherein the primary winding of the power level main transformeris connected with a master switch in series; wherein a firstsemiconductor chip comprises a first controller for producing a firstpulse signal to drive the master switch to be turned on or turned off;wherein a second semiconductor chip comprises a second controller forcomparing a detection voltage representing an output voltage valueand/or a load current value with a first reference voltage, thusdetermining a logic state of a control signal produced thereof accordingto a comparison result; wherein the pulse transformer transfers thelogic state of the control signal from the second controller to thefirst controller, so that the first controller determines the logicstate of the first pulse signal according to the logic state of thecontrol signal, thus determining to turn on or turn off the masterswitch.